MAX16812ATI+T - Maxim Integrated

General Description

The MAX16812 is a peak-current-mode LED driver with

an integrated 0.2Ω power MOSFET designed to control

the current in a single string of high-brightness LEDs

(HB LEDs). The MAX16812 can be used in multiple

converter topologies such as buck, boost, or buck-boost.

The MAX16812 operates over a 5.5V to 76V wide supply

voltage range.

The MAX16812 features a low-frequency, wide-range

brightness adjustment (100:1), analog and PWM dim-

ming control input, as well as a resistor-programmable

EMI suppression circuitry to control the rise and fall times

of the internal switching MOSFET. A high-side LED cur-

rent-sense amplifier and a dimming MOSFET driver are

also included, simplifying the design and reducing the

total component count.

The MAX16812 uses peak-current-mode control, adjust-

able slope compensation that allows for additional design

flexibility. The device has two current regulation loops.

The first loop controls the internal switching MOSFET

peak current, while the second current regulation loop

controls the LED current. Switching frequency can be

adjusted from 125kHz to 500kHz.

Additional features include adjustable UVLO, soft-start,

external enable/disable input, thermal shutdown, a 1.238V

1% accurate buffered reference, and an on-chip oscillator.

An internal 5.2V linear regulator supplies up to 20mA to

power external devices.

The MAX16812 is available in a thermally enhanced 5mm

x 5mm, 28-pin TQFN-EP package and is specified over

the automotive -40°C to +125°C temperature range.

Applications

●Architectural and Industrial Lighting

Features

●Integrated 76V, 0.2Ω (typ) Power MOSFET

●5.5V to 76V Wide Input Range

●Adjustable LED Current with 5% Accuracy

●Floating Differential LED Current-Sense Amplifier

●Floating Dimming N-Channel MOSFET Driver

●PWM LED Dimming with:

• PWM Control Signal

• Analog Control Signal

• Chopped VIN Input

●Peak-Current-Mode Control

●125kHz to 500kHz Adjustable Switching Frequency

●Adjustable UVLO and Soft-Start

●Output Overvoltage Protection

●5µs LED Current Rise/Fall Times During Dimming

Minimize EMI

●Overtemperature and Short-Circuit Protection

19-0880; Rev 1; 4/14

Typical Application Circuit and Pin Configuration appear

at end of data sheet.

+Denotes a lead(Pb)-free/RoHS-compliant package.

*EP = Exposed pad.

PART TEMP RANGE PIN-PACKAGE

MAX16812ATI+ -40°C to +125°C 28 TQFN-EP*

VIN

CIN

RTGRM

CH_REG

ROV1

ROV2

CTGRM

DIM

TGRM

VOUT

L_REG

RCS

DOUT

VOUT

SRC

DRV

SLP

COMP

RCOMP1

BUCK-BOOST CONFIGURATION

CCOMP1

CSLP

RSRC

COUT

RCOMP2

CS-

CS+

DGT

H_REG

SGND

AGND

REFI

REF

CS_OUT

MAX16812

MAX16812 Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

Simplied Diagram

Ordering Information

EVALUATION KIT AVAILABLE

(All voltages are referenced to AGND, unless otherwise noted.)

SGND ...................................................................-0.3V to +0.3V

IN, EN, LX, DIM ..................................................... -0.3V to +80V

L_REG, GT, DRV ....................................................-0.3V to +6V

RT, REF, REFI, CS_OUT, FB, COMP, SRC,

SLP, TGRM, OV ..................................................-0.3V to +6V

LV, HV, CS-, CS+, DGT, DD, H_REG ..................-0.3V to +80V

CS+, DGT, H_REG to LV ......................................-0.3V to +12V

CS- to LV ..............................................................-0.3V to +0.3V

CS+ to CS- ............................................................-0.3V to +12V

DD to LV ...................................................................-1V to +80V

Maximum Current into Any Pin (except LX, SRC) ........... ±20mA

Maximum Current into LX and SRC ......................................+2A

Continuous Power Dissipation (TA = +70°C)

28-Pin TQFN 5mm x 5mm

(derate 34.65mW/°C* above +70°C) .........................2759mW

Operating Temperature Range ......................... -40°C to +125°C

Junction Temperature ...................................................... +150°C

Storage Temperature Range ............................ -65°C to +150°C

Lead Temperature (soldering, 10s) .................................+300°C

(VIN = VEN = 12V, CL_REG = 3.3µF, CH_REG = 1µF, CREF = 47nF, VTGRM = 0V, RSRC = 0.2Ω, TA = TJ = -40°C to +125°C, unless

otherwise noted. Typical values are at TA = +25°C.)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

Input Voltage Range VIN 5.5 76.0 V

Quiescent Supply IQVTGRM = 1V, VDIM = 0V 0.3 2.5 mA

Shutdown Supply Current ISHDN VEN ≤ 300mV 20 45 µA

Internal MOSFET On-Resistance RDSON ILX = 1A, VIN > 10V, VGT = VDRV = 5V 0.2 0.4 Ω

Output Current Accuracy ILED ILED = 350mA, RCS = 1Ω -5 +5 %

Peak Switch Current Limit ILXLIM 2.6 3.1 3.6 A

Hiccup Switch Current 6 A

Switch Leakage Current ILXLEAK VEN = 0V, VLX = 76V, VGT = 0V 1 10 µA

UNDERVOLTAGE LOCKOUT

IN Undervoltage Lockout UVLO VIN rising 4.6 4.9 5.3 V

UVLO Hysteresis 100 mV

EN Threshold Voltage VEN_THUP VEN rising 1.2 1.38 1.6 V

EN Hysteresis 100 mV

REFERENCE (REF) AND LOW-SIDE LINEAR REGULATOR (L_REG)

Startup Response Time tPOR VIN or VEN rising 50 µs

Reference Voltage VREF IREF = 10µA 1.190 1.238 1.288 V

Reference Soft-Start Charging

Current IREF_SLEW VREF = 0V 25 40 60 µA

L_REG Supply Voltage VIN = 7.5V, IL_REG = 1mA 4.9 5.2 5.5 V

L_REG Load Regulation IL_REG = 20mA 20 Ω

L_REG Dropout Voltage IL_REG = 25mA 400 mV

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│

MAX16812 Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

Electrical Characteristics

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these

or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect

device reliability.

*As per JEDEC51 standard (multilayer board).

Absolute Maximum Ratings

(VIN = VEN = 12V, CL_REG = 3.3µF, CH_REG = 1µF, CREF = 47nF, VTGRM = 0V, RSRC = 0.2Ω, RCS = 1Ω, TA = TJ = -40°C to +125°C,

unless otherwise noted. Typical values are at TA = +25°C.)

(VIN = VEN = 12V, CL_REG = 3.3µF, CH_REG = 1µF, CREF = 47nF, VTGRM = 0V, RSRC = 0.2Ω, TA = TJ = -40°C to +125°C, unless

otherwise noted. Typical values are at TA = +25°C.)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

HIGH-SIDE UNDERVOLTAGE LOCKOUT AND LINEAR REGULATOR (H_REG) ((VHV - VLV) = 21V)

H_REG Input-Voltage Threshold VH_REG is rising 3.60 3.887 4.20 V

H_REG Supply Voltage IH_REG = 0 4.75 5 5.40 V

H_REG Load Regulation IH_REG = 0 to 3mA 80 Ω

Dropout Voltage IH_REG = 5mA 820 mV

HIGH-SIDE CURRENT-SENSE AMPLIFIERS (VHV - VLV) = 21V

CS- Input Bias Current ICS- VCS- = VLV, (VCS+ - VCS-) = -0.1V 500 µA

CS+ Input Bias Current ICS+ VCS- = VLV, (VCS+ - VCS-) = 0.1V -1 +1 µA

Input Voltage Range VCS- = VLV 0 0.25 V

Minimum Output Current ICS_OUT

Sinking 25 µA

Sourcing 400

Output Voltage Range VCS_OUT 0 1.5 V

DC Voltage Gain 4 V/V

Unity-Gain Bandwidth 0.8 MHz

Maximum REFI Input Voltage VREFI 1.0 V

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

PWM COMPARATOR

COMP Input Leakage Current ILKCOMP VCOMP = 1V, VSRC = 0.5V, VTGRM = 1V,

VDIM = 0.5V -0.10 +0.10 µA

SRC Input Leakage Current ILKSRC VCOMP = 0V, VSRC = 0.5V, VTGRM = 0V,

VDIM = 0.5V -5 +5 µA

Comparator Offset Voltage VOS(EA) (VCOMP - VSRC) = VOS 860 mV

Input Voltage Range VSRC VCOMP = VSRC + 860mV 0 1.23 V

Propagation Delay tPD 50mV overdrive 100 ns

ERROR AMPLIFIER

FB Input Current VFB = 1V, VREFI = 1.2V -100 +100 nA

REFI Input Current VFB = 1V, VREFI = 1V -100 +100 nA

Error-Amplifier Offset Voltage VOS VFB = VCOMP = 1.2V -23 +23 mV

Input Common-Mode Range VFB = (VCOMP - 0.9V) 0 1.5 V

Source Current ICOMP (VREFI - VFB) ≥ 0.5V 300 µA

Sink Current (VFB - VREFI) ≥ 0.5V 80 µA

COMP Clamp Voltage VCOMP VREF = 1.2V, VFB = 0V 1.20 2.56 V

DC Gain 72 dB

Unity-Gain Bandwidth 0.8 MHz

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│

MAX16812 Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

Electrical Characteristics

Electrical Characteristics (continued)

(VIN = VEN = 12V, CL_REG = 3.3µF, CH_REG = 1µF, CREF = 47nF, VTGRM = 0V, RSRC = 0.2Ω, RCS = 1Ω, TA = TJ = -40°C to +125°C,

unless otherwise noted. Typical values are at TA = +25°C.)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

HIGH-SIDE DIMMING LINEAR REGULATOR ((VHV - VLV) = 21V)

Minimum Output Current IDGT

VLV = VCS-, (VCS+ - VCS-) = 0.3V,

(VDD - VLV) = 1V, VDIM = 1V, VTGRM = 0V,

VDGT = 1V, VREFI = 1.0V, sinking

1.2

VLV = VCS-, (VCS+ - VCS-) = 0.2V,

(VDD - VLV) = 1V, VTGRM = 0V, VDGT = 3V,

VREFI = 1.0V, VDIM = 1V, sourcing

1.2

Output Voltage Range 0.2 5.0 V

DC Gain CDGT = 1nF to LV 60 dB

DD Input Bias Current IDD (VDD - VCS-) = 0.5V -3 +3 µA

DD Input Low Threshold VTGRM = 0V, VDIM = 1V, VREFI = 1.2V,

(VDGT - VLV) > 1.5V, VDD falling 0.25 0.50 0.75 V

DIMMING ((VHV - VLV) = 21V)

DIM Input Bias Current IDIM VDIM = 1.1V -1 +1 µA

TGRM Input High Threshold 1.18 1.23 1.27 V

TGRM Reset High-to-TGRM Low

Pulse Width 1 µs

TGRM Reset Switch RDS(ON) VTGRM = 1.3V 20 Ω

Dimming Rise and Fall LED

Current Times 5 µs

OVERVOLTAGE PROTECTION (OV)

OV Input High Threshold VOV rising 1.180 1.230 1.292 V

OV Input Threshold Hysteresis 14 mV

OV Input Bias Current IOV VOV = 1.1V -1 +1 µA

INTERNAL OSCILLATOR CLOCK

Internal Clock Frequency fOSC

RT = 2MΩ to AGND 470 525 570 kHz

RT = 50kΩ to AGND 105 125 155

SLOPE COMPENSATION INPUT (SLP)

SLP Input Current ISLP VSLP = 0V 150 µA

LOW-SIDE GATE DRIVE (DRV)

DRV Output Low Impedance RDRV_LO DRV sinking 20mA 3 30 Ω

DRV Output High Impedance RDRV_HI DRV sourcing 20mA 10 45 Ω

INTERNAL POWER MOSFET

GT Input Leakage Current VGT = 0 to 5V -1 +1 µA

Internal MOSFET Gate-to-

Source Threshold Voltage VTH 2.5 V

Internal MOSFET Gate Charge QgVLX = 50V 8 nC

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MAX16812 Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

Electrical Characteristics (continued)

(VIN = VEN = 12V, CL_REG = 3.3µF, CH_REG = 1µF, VTGRM = 0V, TA = +25°C, unless otherwise noted.)

EN UVLO THRESHOLD

vs. TEMPERATURE

MAX16812 toc08

TEMPERATURE (°C)

EN UVLO (V)

1109580655035205-10-25

1.05

1.20

1.35

1.50

1.45

1.40

1.30

1.25

1.15

1.10

1.00

-40 125

VEN RISING

IN UVLO THRESHOLD

vs. TEMPERATURE

MAX16812 toc07

TEMPERATURE (°C)

IN UVLO (V)

1109580655035205-10-25

5.01

5.04

5.07

5.10

5.09

5.08

5.06

5.05

5.03

5.02

5.00

-40 125

VIN FALLING

IN UVLO THRESHOLD

vs. TEMPERATURE

MAX16812 toc06

TEMPERATURE (°C)

IN UVLO THRESHOLD (V)

1109580655035205-10-25

5.05

5.10

5.15

5.20

5.00

-40 125

VIN RISING

VREF vs. TEMPERATURE

MAX16812 toc05

TEMPERATURE (°C)

VREF (V)

1109580655035205-10-25

1.22

1.23

1.24

1.25

1.21

-40 125

IREF = 10µA

SHUTDOWN CURRENT

vs. TEMPERATURE

MAX16812 toc04

TEMPERATURE (°C)

SHUTDOWN CURRENT (µA)

1109580655035205-10-25

-40 125

SWITCH CURRENT LIMIT

vs. TEMPERATURE

MAX16812 toc03

TEMPERATURE (°C)

SWITCH CURRENT LIMIT (A)

11095-25 -10 5 35 50 6520 80

2.950

3.000

3.050

3.100

3.150

3.200

3.250

3.300

2.900

-40 125

RDS(ON) vs. VGT

MAX16812 toc02

VGT (V)

RDS(ON) (Ω)

6.45.84.6 5.23.4 4.02.8

0.2

0.4

0.6

0.8

1.0

1.2

1.4

1.6

1.8

2.0

2.2 7.0

TA = +25C

RDS(ON) vs. ILX

MAX16812 toc01

ILX (A)

RDS(ON) (Ω)

2.52.01.5

0.05

0.10

0.15

0.20

0.25

0.30

0.35

0.40

0.45

1.0 3.0

TA = +125C

TA = +25C

TA = -40C

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MAX16812 Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

Typical Operating Characteristics

(VIN = VEN = 12V, CL_REG = 3.3µF, CH_REG = 1µF, VTGRM = 0V, TA = +25°C, unless otherwise noted.)

VH_REG vs. TEMPERATURE

MAX16812 toc15

TEMPERATURE (°C)

VH_REG (V)

1109580655035205-10-25

4.3

4.6

4.9

5.2

5.1

5.0

4.8

4.7

4.5

4.4

4.2

-40 125

(VHV - VLV) = 21V

ILOAD = 3mA

VH_REG vs. IH_REG

MAX16812 toc14

IH_REG (mA)

VH_REG (V)

2.52.01.51.00.5

4.55

4.60

4.65

4.70

4.75

4.80

4.85

4.90

4.95

5.00

4.50

0 3.0

(VHV - VLV) = 6V

VIN = 12V

VH_REG IS MEASURED

WITH RESPECT TO VLV

VH_REG THRESHOLD

vs. TEMPERATURE

MAX16812 toc13

TEMPERATURE (°C)

VH_REG THRESHOLD (V)

1109580655035205-10-25

3.6

3.9

4.2

4.1

4.0

3.8

3.7

3.5

3.4

-40 125

OSCILLATOR FREQUENCY vs. RT

MAX16812 toc12

RT (MΩ)

OSCILLATOR FREQUENCY (kHz)

10.1

100

200

300

400

500

600

0.01 10

OSCILLATOR FREQUENCY

vs. TEMPERATURE

MAX16812 toc11

TEMPERATURE (°C)

OSCILLATOR FREQUENCY (kHz)

1109580655035205-10-25

100

200

300

400

500

600

-40 125

RT = 2MΩ

RT = 180kΩ

RT = 50kΩ

VL_REG vs. IL_REG

MAX16812 toc10

IL_REG (mA)

VL_REG (V)

181612 144 6 8 102

4.6

4.7

4.8

4.9

5.0

5.1

5.2

5.3

5.4

5.5

4.5

0 20

TA = +125C

TA = +25C

TA = -40C

VIN = 7.5V

EN UVLO THRESHOLD

vs. TEMPERATURE

MAX16812 toc09

TEMPERATURE (°C)

EN UVLO (V)

1109580655035205-10-25

1.05

1.20

1.35

1.50

1.45

1.40

1.30

1.25

1.15

1.10

1.00

-40 125

VEN FALLING

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MAX16812 Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

Typical Operating Characteristics (continued)

PIN NAME FUNCTION

1 FB Low-Side Error Amplifier’s Inverting Input

2 COMP Low-Side Error Amplifier’s Output. Connect a compensation network from COMP to FB for stable operation.

3 REFI Reference Input. VREFI provides the reference voltage for the high-side current-sense amplifier to set the

LED current.

4 REF +1.23V Reference Output. Connect an appropriate soft-start capacitor from REF to AGND.

5 CS_OUT High-Side Current-Sense Amplifier Output. VCS_OUT is proportional to the current through RCS.

6 AGND Analog Ground

7 EN

Enable Input/Undervoltage Lockout. Connect EN to IN through a resistive voltage-divider to program the

UVLO threshold. Connect EN directly to IN to set up the device for 5V internal threshold. Apply a logic-level

input to EN to enable/disable the device.

8 IN Positive Power-Supply Input. Bypass with a 1µF ceramic capacitor to AGND.

9 L_REG 5V Low-Side Regulator Output. Bypass with a 3.3µF ceramic capacitor to AGND.

10 SGND Signal Ground

11 DD MOSFET’s Drain Voltage-Sense Input. Connect DD to the drain of the external dimming MOSFET.

12 DGT External Dimming MOSFET’s Gate Drive

13 CS+ High-Side Current-Sense Amplifier’s Positive Input. Connect RCS between CS+ and CS-. CS+ is

referenced to LV.

14 CS- High-Side Current-Sense Amplifier’s Negative Input. Connect RCS between CS- and CS+. CS- is

referenced to LV.

15 LV High-Side Reference Voltage Input. A DC voltage at LV sets the lowest reference point for the high-side

current-sense and dimming MOSFET control circuitry.

16 H_REG High-Side Regulator Output. H_REG provides a regulated supply for high-side circuitry. Bypass with a 1µF

ceramic capacitor to LV.

17 HV High-Side Positive Supply Voltage Input. HV provides power for dimming and LED current-sense circuitry.

HV is referenced to LV.

18 DRV Internal MOSFET Gate Driver Output. Connect to a resistor between DRV and GT to set the rise and fall

times at LX.

19 GT Internal MOSFET GATE. Connect a resistor between GT and DRV to set the rise and fall times at LX.

20, 21 LX Internal MOSFET Drain

22, 23 SRC Internal Power MOSFET Source

24 SLP Slope Compensation Setting. Connect an appropriate external capacitor from SLP to AGND to generate a

ramp signal for stable operation.

25 TGRM Dimming Comparator’s Reference/Ramp Generator

26 DIM Dimming Control Input

27 RT Resistor-Programmable Internal Oscillator Setting. Connect a resistor from RT to AGND to set the internal

oscillator frequency.

28 OV

Overvoltage Protection Input. Connect OV to HI through a resistive voltage-divider to AGND to set the

overvoltage limit for the load. When the voltage at OV exceeds the 1.238V (typ) threshold, the gate drive

(DRV) for the switching MOSFET is disabled. Once VOV goes below 1.238V by 14mV, the switching

MOSFET turns on again.

— EP Exposed Pad. Connect EP to a large-area ground plane for effective power dissipation. Do not use as the

IC ground connection.

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MAX16812 Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

Pin Description

Figure 1. Functional Diagram

LDOH

POR

3.88V 1.2X

1.1X

PREG BG

VREF

LDOL

OSC

tD = 200ns

UVLO/

POR

DIM

RAMP

REF

CMP

IHI CSA

ADIM

DRMP

0.5V

LATCH

LOGIC

CONTROL

ERROR

AMPLIFIER

AND

DIMMING

S/H

VREFI = 1.2V

VRAMP = 0.3V

CMP

OVP

CMP

1.238V

2µs PULSE

LOW TO DISCHARGE

DIM

SIGNAL

2.5V

VDD

1.2V

SGND

HICCUP

0.6V

ILIM

PWM

VBE

X0.2

MAX16812

H_REG

L_REG

REF

DIM

TGRM

SGND AGND

CS+

CS-

SRC

DRV

SLP

COMP

CS_OUT

REFI

DGT

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MAX16812 Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

Detailed Description

The MAX16812 is a current-mode PWM LED driver with

an integrated 0.2Ω power MOSFET for use in driving HB

LEDs. By using two current regulation loops, 5% LED

current accuracy is achieved. One current regulation

loop controls the internal MOSFET peak current through

a sense resistor (RSRC) from SRC to ground, while the

other current regulation loop controls the average LED

current in a single LED string through another sense

resistor (RCS) in series with the LEDs.

The MAX16812 includes a cycle-by-cycle current limit

that turns off the gate drive to the internal MOSFET

during an overcurrent condition. The MAX16812 features

a programmable oscillator that simplifies and optimizes

the design of magnetics. The MAX16812 is well suited

for inputs from 5.5V to 76V. An external resistor in series

with the internal MOSFET gate can control the rise and

fall times on the drain of the internal switching MOSFET,

therefore minimizing EMI problems.

The MAX16812 high-frequency, current-mode PWM

HB LED driver integrates all the necessary building blocks

for driving a series LED string in an adjustable constant

current mode with PWM dimming. Current-mode control

with leading-edge blanking simplifies control-loop design,

and an external adjustable slope-compensation control

stabilizes the inner current-mode loop when operating at

duty cycles above 50%.

An input undervoltage lockout (UVLO) programs the input

supply startup voltage. An external voltage-divider on

EN programs the supply startup voltage. If EN is directly

connected to the input, the UVLO is set at 5V. A single

external resistor from RT to AGND programs the switch-

ing frequency from 125kHz to 500kHz.

Wide contrast (100:1) PWM dimming can be achieved

with the MAX16812. A DC input on DIM controls the

dimming duty cycle. The dimming frequency is set by

the sawtooth ramp frequency on TGRM (see the PWM

Dimming section). In addition, PWM dimming can be

achieved by applying a PWM signal to DIM with TGRM

set to a DC voltage less than 1.238V. A floating high-volt-

age driver drives an external n-channel MOSFET in series

with the LED string. REFI allows analog dimming of the

LED current, further increasing the effective dimming

range over PWM alone. The MAX16812 has a 5µs pre-

programmed LED current rise and fall time.

A nonlatching overvoltage protection limits the voltage on

the internal switching MOSFET under open-circuit condi-

tions in the LED string. The internal thermal shutdown cir-

cuit protects the device if the junction temperature should

exceed +165°C.

Current-Mode Control

The MAX16812 offers a current-mode control opera-

tion feature with leading-edge blanking that blanks the

sensed current signal applied to the input of the PWM

current-mode comparator. In addition, a current-limit com-

parator monitors the same signal at all times and provides

cycle-by-cycle current limit. An additional hiccup com-

parator limits the absolute peak current to two times the

cycle-by-cycle current limit. The leading-edge blanking of

the current-sense signal prevents noise at the PWM com-

parator input from prematurely terminating the on-cycle.

The switch current-sense signal contains a leading-edge

spike that results from the MOSFET gate-charge current,

and the capacitive and diode reverse-recovery current of

the power circuit. The MAX16812’s capacitor-adjustable

slope-compensation feature allows for easy stabilization

of the inner switching MOSFET current-mode loop. Upon

triggering the hiccup current limit, the soft-start capacitor

on REF is discharged and the gate drive to DRV is dis-

abled. Once the inductor current falls below the hiccup

current limit, the soft-start capacitor is released and it

begins to charge after 10µs.

Slope Compensation

The MAX16812 uses an internal ramp generator for

slope compensation. The internal ramp signal resets at

the beginning of each cycle and slews at the rate pro-

grammed by the external capacitor connected at SLP

and an internal ISLP current source of 150µA. An internal

attenuator attenuates the actual slope compensation

signal by a factor of 0.2. Adjust the MAX16812 slew-rate

capacitor by using the following equation:

SLP

SLOPE

C 0.2 SR

= ×

where ISLP is the charging current in mA and CSLOPE is

the slope compensation capacitance on the SLP in µF,

and SR is the designed slope in mV/µs.

When using the MAX16812 for internal switching MOSFET

duty cycles greater than 50%, the following conditions

must be met to avoid current-loop subharmonic oscilla-

tions.

SRC IND_OFF

0.5 R V

SR mV / s

××

≥µ

where RSRC is in mΩ, VIND_OFF is in volts, and L is in

µH. L is the inductor connected to the LX pin of the

internal switching MOSFET and VIND_OFF is the voltage

across the inductor during the off-time of the internal

MOSFET.

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MAX16812 Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

Undervoltage Lockout

The MAX16812 features an adjustable UVLO through the

enable input (EN). Connect EN directly to IN to use the 5V

default UVLO. Connect EN to IN through a resistive divid-

er to ground to set the UVLO threshold. The MAX16812

is enabled when VEN exceeds the 1.38V (typ) threshold.

Calculate the EN UVLO resistor-divider values as follows

(see Figure 2):

UV1 UV2

UVLO EN

R R x

V - V







where RUV1 is in the 20kΩ range, VEN is the 1.38V

(typ) EN threshold voltage, and VUVLO is the desired

input-voltage UVLO threshold in volts. Due to the 100mV

hysteresis of the UVLO threshold, capacitor CEN is

required to prevent chattering at the UVLO threshold due

to line impedance drops at power-up and during dimming.

If the undervoltage setting is very close to the required

minimum operating voltage, there can be jumps in the

voltage at IN while dimming. CEN should be large enough

to limit the ripple on EN to less than 100mV (EN hystere-

sis) under these conditions so that it does not turn on and

off due to the ripple on IN.

Soft-Start

The soft-start feature of the MAX16812 allows the LED

string current to ramp up in a controlled manner, thus

minimizing output-voltage overshoot. While the part is in

UVLO, CREF is discharged (Figure 3). Upon coming out

of UVLO, an internal current source starts charging CREF

during the soft-start cycle. Use the following equation to

calculate total soft-start e:

ST REF REF

1.238

t C I

= ×

where IREF is 40µA, CREF is in µF, and tST is in seconds.

Operation begins when REF ramps above 0.6V. Once

the soft-start is complete, REF is regulated to 1.238V, the

internal voltage reference.

Low-Side Internal

Switching MOSFET Driver Supply (L_REG)

L_REG is the regulated (5.2V) internal supply voltage

capable of delivering 20mA. L_REG provides power to

the gate drive of the internal switching power MOSFET.

VL_REG is referenced to AGND. Connect a 3.3µF ceramic

capacitor from L_REG to AGND.

High-Side Regulator (H_REG)

H_REG is a low-dropout linear regulator referenced

to LV. H_REG provides the gate drive for the external

n-channel dimming MOSFET and also powers up the

MAX16812’s LED current-sense circuitry. Bypass H_REG

to LV with a 1µF ceramic capacitor.

Figure 3. Soft-Start SettingFigure 2. UVLO Threshold Setting

MAX16812

VIN

AGND

REF

CREF

MAX16812

VIN

AGND

RUV2

RUV1

CEN

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MAX16812 Integrated High-Voltage LED Driver

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High-Side Current-Sense Output (CS_OUT)

A high-side transconductance amplifier converts the volt-

age across the LED current-sense resistor (RCS) into

an internal current output. This current flows through an

internal resistor connected to AGND. The voltage gain for

the LED current-sense signal is 4. The amplified signal is

then buffered and connected through an internal switch

to CS_OUT.

Internal Error Amplier

The MAX16812 includes a built-in voltage-error amplifi-

er, which can be used to close the feedback loop. The

internal LED current-sense output signal is buffered

internally and then connected to CS_OUT through an

internal switch. CS_OUT is connected to the inverting

input (FB) pin of the error amplifier through a resistor.

See Figures 4 and 5. The reference voltage for the out-

put current is connected to REFI, the noninverting input

of the error amplifier. When the internal dimming signal

is low, COMP is disconnected from the output of the error

amplifier and CS_OUT is simultaneously disconnected

from the buffered LED current-sense output signal (Figure

5). When the internal dimming signal is high, the output

of the op amp is connected to COMP and CS_OUT is

connected to the buffered LED current-sense signal at

the same time (Figure 4). This enables the compensation

capacitor to hold the charge when the DIM signal has

turned off the internal switching MOSFET gate drive.

To maintain the charge on the compensation capacitors

CCOMP1 and CCOMP2, the capacitors should be of the

low-leakage ceramic type.

When the internal dimming signal is enabled, the volt-

age on the compensation capacitor forces the converter

into steady state almost instantaneously. The voltage on

COMP is subtracted from the internal slope compensation

signal and is then connected to one of the inputs of the

PWM comparator. The PWM comparator input is of the

CMOS type with very low bias currents.

Figure 5. Internal Error Amplifier Connections (Dimming Signal Low)

Figure 4. Internal Error Amplifier Connection (Dimming Signal High)

REFI

COMP

OUT

STATE B

RCOMP2

RCOMP1

CCOMP1

CCOMP2

REFI

COMP

OUT

RCOMP2

RCOMP1

CCOMP1

CCOMP2

STATE A

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MAX16812 Integrated High-Voltage LED Driver

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Analog Dimming

The MAX16812 offers analog dimming of the LED current

by allowing the application of an external voltage at REFI.

The output current is proportional to the voltage at REFI.

Use a potentiometer from REF or directly apply an exter-

nal voltage source at REFI.

PWM Comparator

The PWM comparator uses the instantaneous switch

current, the error-amplifier output, and the slope com-

pensation to determine when the gate drive DRV to the

internal n-channel switching MOSFET turns off. In normal

operation, gate drive DRV to the n-channel MOSFET

turns off when:

ISW x RSRC ≥ VCOMP - VOFFSET - VSCOMP

where ISW is the current through the internal n-channel

switching MOSFET, RSRC is the switch current-sense

resistor, VCOMP is the output voltage of the internal ampli-

fier, VOFFSET is the internal DC offset, which is a VBE

drop, and VSCOMP is the ramp function that starts at zero

and slews at the programmed slew rate (SR).

Internal Switching MOSFET Current Limit

The current-sense resistor (RSRC), connected between

the source of the internal MOSFET and ground, sets

the current limit. The SRC input has a voltage trip level

(VSRC) of 600mV for the cycle-by-cycle current limit. Use

the following equation to calculate the value RSRC:

SRC

SRC LXLIM

where ILXLIM is the peak current that flows through the

switching MOSFET at full load and low line. When the

voltage produced by this current (through the cur-

rent-sense resistor) exceeds the current-limit (ILIM) com-

parator threshold, the MOSFET driver (DRV) quickly

terminates the current on-cycle. The 200ns leading-edge

blanking circuit suppresses the leading-edge spike on

the current-sense waveform from appearing at the cur-

rent-limit comparator. There is also a hiccup comparator

(HICCUP) that limits the peak current in the internal

switch set at twice the peak limit setting.

Internal n-Channel

Switching MOSFET Driver (DRV)

L_REG provides power for the DRV output. Connect a

resistor from DRV to gate GT of the internal switching

MOSFET to control the switching MOSFET rise and fall

times, if necessary.

External Dimming

MOSFET Gate Drive (DGT)

DGT is the gate drive to the external dimming MOSFET

referenced to LV. H_REG provides the power to the gate

drive.

Overvoltage Protection

The overvoltage protection (OVP) comparator compares

the voltage at OV with a 1.238V (typ) internal reference.

When the voltage at OV exceeds the internal reference,

the OVP comparator terminates PWM switching and no

further energy is transferred to the load. Connect OV to

HV through a resistive voltage-divider to ground to set the

overvoltage threshold at the output.

Setting the Overvoltage Threshold

Connect OV to HV or to the high-side of the LEDs through

a resistive voltage-divider to set the overvoltage threshold

at the output (Figure 6).

Figure 6. OVP Setting

MAX16812

VLED+

AGND

ROV1

ROV2

MAX16812

VLED+

AGND

ROV1

ROV2

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MAX16812 Integrated High-Voltage LED Driver

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The overvoltage protection (OVP) comparator compares

the voltage at OV with a 1.238V (typ) internal reference.

Use the following equation to calculate resistorlues:

OV_LIM OV

OV1 OV2 OV

V V

R R x

−







where VOV is the 1.238V OV threshold. Choose ROV1

and ROV2 to be reasonably high-value resistors to pre-

vent the discharge of filter capacitors. This prevents

degraded performance during dimming.

Internal Oscillator Switching Frequency

The oscillator switching frequency is programmed by a

resistor connected from RT to AGND. To program the

oscillator frequency above 125kHz, choose the appropri-

ate resistor RT from the curves shown in the Oscillator

Frequency vs. RT graph in the Typical Operating

Characteristics section.

PWM Dimming

PWM dimming can be achieved by driving DIM with an

analog voltage less than VREF. See Figure 7. An external

resistor on TGRM from L_REG in conjunction with the

ramp capacitor, CTGRM, from TGRM to AGND creates a

sawtooth ramp that is compared with the DC voltage on

DIM. The output of the comparator is a pulsating dimming

signal. The frequency fRAMP of the sawtooth signal on

TGRM is given by:

RAMP

TGRM TGRM

3.67

f C R

≅×

Use the following formula to calculate the voltage VDIM,

necessary for a given output duty cycle, D:

VDIM = D x 1.238V

where VDIM is the DC voltage applied to DIM in volts.

The DC voltage for DIM can also be created by connect-

ing DIM to REF through a resistive voltage-divider. Using

the required dimming input voltage, VDIM, calculate the

resistor values for the divider string using the following

equation:

RDIM2 = [VDIM / (VREF - VDIM)] x RDIM1

where VREF is the voltage on REF.

PWM dimming can also be achieved by connecting

TGRM to a DC voltage less than VREF and applying the

PWM signal at DIM. The moment the internal dimming

signal goes low, gate drive DRV to the internal switching

MOSFET is turned off. The error amplifier goes to state

B (see the Internal Error Amplifier section and Figures 4

and 5). The peak current in the inductor prior to disabling

DRV is ILX. Gate drive DGT to the external dimming

MOSFET is held high. Then after a switchover period,

gate voltage VDGT on the external dimming MOSFET is

linearly controlled to reduce the LED current to 0. The fall

time of the LED current is controlled by an internal timing

circuit to 5µs for the MAX16812. During this period, the

gate (DRV) to the internal switching MOSFET is enabled.

After the fall time, the gate drive to the external dimming

MOSFET is turned off and the gate drive to the internal

switching MOSFET is still held high after the switchover

period. The peak current in the inductor is controlled at

ILX. Then after a time period of 20µs, the gate drive is

disabled. The scope shots in Figures 8–11 show the dim-

ming waveforms.

Figure 7. PWM Dimming from REF

MAX16812

AGND

DIM

REF

TGRM

L_REG

RDIM1 RTGRM

CTGRM

RDIM2

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MAX16812 Integrated High-Voltage LED Driver

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When the DIM signal goes high, the LED current is grad-

ually increased to the programmed value. The rise time

of the LED current is controlled to 5µs for the MAX16812

by controlling the voltage on DGT. After the rise time, an

internal sensing circuit monitors the voltage across the

drain to the source of the external dimming MOSFET. The

LED current is now controlled at the programmed value

by a linear current regulating circuit. Once the voltage

across the drain to source of the dimming MOSFET drops

below 0.5V, the reference for the linear current regulating

circuit is increased to 1.1 times the programmed value.

The gate drive (DRV) to the internal switching MOSFET is

enabled and the error amplifier is returned to state A (see

the Internal Error Amplifier section and Figures 4 and 5).

Fault Protection

The MAX16812 features built-in overvoltage protection

and thermal shutdown. Connect a resistive voltage-di-

vider between HV, OV, and AGND to program the over-

voltage protection. In the case of a short circuit across

the LED string, the temperature of the external dimming

MOSFET could exceed the maximum allowable junction

temperature. This is due to excess power dissipation in the

MOSFET. Use the fault protection circuit shown in Figure

12 to protect the external dimming MOSFET.

Internal thermal shutdown in the MAX16812 safely turns

off the IC when the junction temperature exceeds +165°C.

Figure 11. LED Current, DIM Signal, and DRV Waveforms

when DIM Signal Goes High

Figure 10. LED Current, Output Voltage, and DRV Waveforms

when DIM Signal Goes High

Figure 9. LED Current, DIM Signal, and DRV Waveforms when

DIM Signal Goes Low

Figure 8. LED Current, Output Voltage, and DRV Waveforms

when DIM Signal Goes Low

MAX16812 fig11

10s/div

VDRV

2V/div

0A, 0V

100mA/div

VDIM

5V/div

ILED

MAX16812 fig10

10s/div

ILED

VOUT

VDRV 0V

2V/div

0A, 0V

100mA/div

10V/div

MAX16812 fig09

10s/div

VDIM

5V/div

VDRV

2V/div

0A, 0V

ILED

100mA/div

MAX16812 fig08

10s/div

ILED

VOUT

VDRV 0V

2V/div

0A, 0V

100mA/div

10V/div

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MAX16812 Integrated High-Voltage LED Driver

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Inductor Selection

The minimum required inductance is a function of the

operating frequency, the input-to-output voltage differen-

tial and the peak-to-peak inductor current (∆IL). Higher

∆IL allows for a lower inductor value while a lower ∆IL

requires a higher inductor value. A lower inductor value

minimizes size and cost, improves large-signal transient

response, but reduces efficiency due to higher peak

currents and higher peak-to-peak output ripple voltage

for the same output capacitor. On the other hand, higher

inductance increases efficiency by reducing the ripple

current, ∆IL. However, resistive losses due to the extra

turns can exceed the benefit gained from lower ripple cur-

rent levels, especially when the inductance is increased

without allowing for larger inductor dimensions. A good

compromise is to choose ∆IL equal to 30% of the full

load current. The inductor saturating current specification

is also important to avoid runaway current during output

overload and continuous short-circuit conditions.

Buck Configuration: In a buck configuration (Figure 13),

the average inductor current does not vary with the input.

The worst-case peak current occurs at the highest input

voltage. In this case, the inductance, L, for continuous

conduction mode given by:

( )

OUT INMAX OUT

INMAX SW L

V x V V

L V x f x I

−

=∆

where VINMAX is the maximum input voltage, fSW is the

switching frequency, and VOUT is the output voltage.

Boost Configuration: In the boost converter, the aver-

age inductor current varies with the input voltage and the

maximum average current occurs at the lowest input volt-

age. For the boost converter, the average inductor current

is equal to the input current. In this case, the inductance,

L, is calculated as:

( )

INMIN OUT INMIN

OUT SW L

V x V V

L V x f x I

−

=∆

where VINMIN is the minimum input voltage, VOUT is the

output voltage, and fSW is the switching frequency. See

Figure 14.

Buck-Boost Configuration: In a buck-boost converter

(see the Typical Application Circuit), the average inductor

current is equal to the sum of the input current and the

LED current. In this case, the inductance, L, is:

( )

OUT INMIN

OUT INMIN SW L

V x V

V V x f x I

+∆

where VINMIN is the minimum input voltage, VOUT is the

output voltage, and fSW is the switching frequency.

Figure 12. Dimming MOSFET Protection

MAX6501

GND

4.7F

100k

5.1V

ZENER

GND

VCC

TO L_REG PIN

OF MAX16812

TO EN PIN OF

MAX16812

VIN

TOVER

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MAX16812 Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

Figure 14. Boost Configuration

Figure 13. Buck Configuration

MAX16812

LV SRC

TGRM

L_REG

CS- CS+ DGT DD H_REG HV LX

RCS

RRT

RTGRM

CH_REG

CL_REG

CREF

ROV1

RCOMP1

RREF1

RREF2

RCOMP2

CCOMP1

CCOMP2

VOUT

VIN

ROV2

DRV

SLP

COMP

OV SGND AGND REFI

REF FBCS_OUT

VIN

CIN1

COUT

DIM

CTGRM

CSLP

DOUT

RSRC

VOUT

MAX16812

H_REG

TGRM

L_REG

HV LX LV DD DGT CS- CS+

RRT

RTGRM

CL_REG

CH_REG

CREF

ROV1

RCOMP1

RREF1

RREF2

RCOMP2

CCOMP1

CCOMP2

VOUT

ROV2

DRV

SLP

COMP

OV SGND AGND REFI

REF FBCS_OUT

VIN

DIM

CTGRM

CIN

CSLP

SRC

RSRC

DOUT RCS

COUT

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MAX16812 Integrated High-Voltage LED Driver

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Output Capacitor

The function of the output capacitor is to reduce the out-

put ripple to acceptable levels. The ESR, ESL, and the

bulk capacitance of the output capacitor contribute to the

output ripple. In most of the applications, the output ESR

and ESL effects can be dramatically reduced by using

low-ESR ceramic capacitors. To reduce the ESL effects,

connect multiple ceramic capacitors in parallel to achieve

the required capacitance.

In a buck configuration, the output capacitance, COUT, is

calculated using the follow equation:

INMAX OUT OUT

OUT 2

R INMAX SW

( V V ) V

V 2 L V f

−×

≥∆ ×× × ×

where ∆VR is the maximum allowable output ripple.

In a boost configuration, the output capacitance, COUT,

is calculated as:

OUT INMIN OUT

OUT R OUT SW

( V V ) 2 I

C V V f

− ××

≥∆× ×

where COUT is the output capacitor.

In a buck-boost configuration, the output capacitance,

COUT is:

OUT OUT

OUT

R OUT INMIN SW

2 V I

V (V V ) f

××

≥∆× + ×

where VOUT is the voltage across the load and IOUT is

the output current.

Input Capacitor

An input capacitor connected between IN and ground

must be used when configuring the MAX16812 as a buck

converter. Use a low-ESR input capacitor that can handle

the maximum input RMS ripple current. Calculate the

maximum RMS ripple using the follow equation:

OUT OUT INMIN OUT

IN(RMS)

INMIN

I V (V - V )

××

When using the MAX16812 in a boost or buck-boost con-

figuration, the input capacitor’s RMS current is low and

the input capacitance can be small. However, an addi-

tional electrolytic capacitor may be required to prevent

oscillations due to line impedances.

Figure 15. SEPIC Configuration

MAX16812

CS-

CS+

DGT

H_REG

SGND

AGND

REFI

REF

CS_OUT

RREF2

CH_REG

RREF1

CSLP

COUT

RCOMP1

RCOMP2

ROV1

ROV2

RSRC

VOUT

CL_REG

RCS

RTGRM

COMP

SLP

SRC

DRV

CCOMP1

CCOMP2

DOUT

VOUT

VIN

CIN1

CTGRM

VIN

L_REG

TGRM

DIM

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MAX16812 Integrated High-Voltage LED Driver

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Layout Recommendations

Typically, there are two sources of noise emission in a

switching power supply: high di/dt loops and high dv/dt

surfaces. For example, traces that carry the drain current

often form high di/dt loops. Similarly, the drain of the

internal MOSFET connected to the LX pin presents a dv/

dt source. Keep all PCB traces carrying switching cur-

rents as short as possible to minimize current loops. Use

ground planes for best results.

Careful PCB layout is critical to achieve low switching

losses and clean, stable operation. Use a multilayer board

whenever possible for better noise immunity and power

dissipation. Follow these guidelines for good PCB layout:

●Use a large copper plane under the MAX16812 pack-

age. Ensure that all heat-dissipating components

have adequate cooling. Connect the exposed pad of

the device to the ground plane.

●Isolate the power components and high-current paths

from sensitive analog circuitry.

●Keep the high-current paths short, especially at the

ground terminals. This practice is essential for stable,

jitter-free operation. Keep switching loops short.

●Connect AGND and SGND to a ground plane.

Ensure a low-impedance connection between all

ground points.

●Keep the power traces and load connections short.

This practice is essential for high efficiency. Use thick

copper PCBs to enhance full-load efficiency.

●Ensure that the feedback connection to FB is short

and direct.

●Route high-speed switching nodes away from the

sensitive analog areas.

●To prevent discharge of the compensation capacitors,

CCOMP1 and CCOMP2, during the off-time of the dim-

ming cycle, ensure that the PCB area close to these

components has extremely low leakage.

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MAX16812 Integrated High-Voltage LED Driver

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PACKAGE

TYPE

PACKAGE

CODE

OUTLINE

NO.

LAND

PATTERN NO.

28 TQFN-EP T2855+8 21-0140 90-0028

MAX16812

TQFN

TOP VIEW

COMP

REF

CS_OUT

AGND

DRV

H_REG

1 2

TGRM

4 5 6 7

2021 19 17 16 15

DIM

DGT

SGND

L_REG

REFI GT

28 8

OV IN

*EP

*EP = EXPOSED PAD

SLP

23 13 CS+

SRC

22 14 CS-

SRC

MAX16812

BUCK-BOOST CONFIGURATION

SRC

TGRM

L_REG

CS- CS+ DGT DD H_REG HV LX

RCS

RTGRM

CH_REG

CL_REG

ROV1

RCOMP1

RREF1

RREF2

CREF

RCOMP2

CCOMP1

CCOMP2

VOUT

ROV2

DRV

SLP

COMP

OV SGND AGND REFI

REF FBCS_OUT

VIN

CIN1

COUT

DIM

CTGRM

CSLP

DOUT

RSRC

VOUT

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MAX16812 Integrated High-Voltage LED Driver

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Chip Information

PROCESS: BiCMOS

TRANSISTOR COUNT: 8699

Pin Conguration

Typical Application Circuit

Package Information

For the latest package outline information and land patterns

(footprints), go to www.maximintegrated.com/packages. Note

that a “+”, “#”, or “-” in the package code indicates RoHS status

only. Package drawings may show a different suffix character, but

the drawing pertains to the package regardless of RoHS status.

REVISION

NUMBER

REVISION

DATE DESCRIPTION PAGES

CHANGED

0 7/07 Initial release —

1 4/14 No /V OPNs; removed Automotive reference from Applications section 1

Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses

are implied. Maxim Integrated reserves the right to change the circuitry and specications without notice at any time. The parametric values (min and max limits)

shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.

Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.

MAX16812 Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

│

Revision History

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