1
LTC1627
Monolithic Synchronous
Step-Down Switching Regulator
Figure 1b. Efficiency vs Output Load Current
Figure 1a. High Efficiency Step-Down Converter
The LTC
®
1627 is a monolithic current mode synchronous
buck regulator using a fixed frequency architecture. The
operating supply range is from 8.5V down to 2.65V, making
it suitable for one or two lithium-ion battery-powered appli-
cations. Burst Mode operation provides high efficiency at
low load currents. 100% duty cycle provides low dropout
operation, which extends operating time in battery-operated
systems.
The operating frequency is internally set at 350kHz, allowing
the use of small surface mount inductors. For switching noise
sensitive applications it can be externally synchronized up to
525kHz. The SYNC/FCB control pin guarantees regulation of
secondary windings regardless of load on the main output by
forcing continuous operation. Burst Mode operation is inhib-
ited during synchronization or when the SYNC/FCB pin is
pulled low to reduce noise and RF interference. Soft-start is
provided by an external capacitor.
Optional bootstrapping enhances the internal switch drive for
single lithium-ion cell applications. The internal synchronous
switch increases efficiency and eliminates the need for an
external Schottky diode, saving components and board
space. The LTC1627 comes in an 8-lead SO package.
L1 15µH
C
OUT
100µF
6.3V
80.6k
249k
47pF
V
IN
2.8V*
TO 8.5V
V
OUT
3.3V
1627 F01a
C
SS
0.1µF
C
IN
22µF
16V
1
2
3
4
8
7
6
5
SYNC/FCB
V
DR
V
IN
SW
I
TH
RUN/SS
V
FB
GND
LTC1627
++
*V
OUT
CONNECTED TO V
IN
FOR 2.8V < V
IN
< 3.3V
OUTPUT CURRENT (mA)
1
70
EFFICIENCY (%)
90
95
100
10 100 1000
1627 F01b
85
80
75
VOUT = 3.3V
VIN = 6V
VIN = 3.6V
VIN = 8.4V
■
High Efficiency: Up to 96%
■
Constant Frequency 350kHz Operation
■
2.65V to 8.5V V
IN
Range
■
V
OUT
from 0.8V to V
IN
, I
OUT
to 500mA
■
No Schottky Diode Required
■
Synchronizable Up to 525kHz
■
Selectable Burst Mode
TM
Operation
■
Low Dropout Operation: 100% Duty Cycle
■
Precision 2.5V Undervoltage Lockout
■
Secondary Winding Regulation
■
Current Mode Operation for Excellent Line and
Load Transient Response
■
Low Quiescent Current: 200µA
■
Shutdown Mode Draws Only 15µA Supply Current
■
±1.5% Reference Accuracy
■
Available in 8-Lead SO Package
■
Cellular Telephones
■
Portable Instruments
■
Wireless Modems
■
RF Communications
■
Distributed Power Systems
■
Scanners
■
Single and Dual Cell Lithium
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a trademark of Linear Technology Corporation.
FEATURES
DESCRIPTIO
U
APPLICATIO S
U
TYPICAL APPLICATIO
U
2
LTC1627
T
JMAX
= 125°C, θ
JA
= 110°C/W
ORDER PART
NUMBER
S8 PART MARKING
TOP VIEW
SYNC/FCB
V
DR
V
IN
SW
I
TH
RUN/SS
V
FB
GND
S8 PACKAGE
8-LEAD PLASTIC SO
1
2
3
4
8
7
6
5
Consult factory for Military grade parts.
(Note 1)
Input Supply Voltage ................................ –0.3V to 10V
Driver Supply Voltage (V
IN
– V
DR
) ........... –0.3V to 10V
I
TH
Voltage .................................................. –0.3V to 5V
Run/SS, V
FB
Voltages ................................ –0.3V to V
IN
Sync/FCB Voltage ...................................... –0.3V to V
IN
V
DR
Voltage (V
IN
≤ 5V) ............................... –5V to 0.3V
P-Channel Switch Source Current (DC) .............. 800mA
N-Channel Switch Sink Current (DC) .................. 800mA
Peak SW Sink and Source Current ......................... 1.5A
Operating Ambient Temperature Range
Commercial ............................................ 0°C to 70°C
Industrial ........................................... –40°C to 85°C
Junction Temperature (Note 2).............................125°C
Storage Temperature Range ................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec)..................300°C
1627
1627I
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
I
VFB
Feedback Current (Note 3) 20 60 nA
V
FB
Regulated Feedback Voltage (Note 3)
●
0.788 0.80 0.812 V
∆V
OVL
∆Output Overvoltage Lockout ∆V
OVL
= V
OVL
– V
FB
20 60 110 mV
∆V
FB
Reference Voltage Line Regulation V
IN
= 2.8V to 8.5V (Note 3) 0.002 0.01 %/V
V
LOADREG
Output Voltage Load Regulation I
TH
Sinking 2µA (Note 3) 0.5 0.8 %
I
TH
Sourcing 2µA (Note 3) –0.5 –0.8 %
I
S
Input DC Bias Current (Note 4)
Synchronized V
IN
= 8.5V, V
OUT
= 3.3V, Frequency = 525kHz 450 µA
Burst Mode Operation V
ITH
= 0V, V
IN
= 8.5V, V
SYNC/FCB
= Open 200 320 µA
Shutdown V
RUN/SS
= 0V, 2.65V < V
IN
< 8.5V 15 35 µA
Shutdown V
RUN/SS
= 0V, V
IN
< 2.65V 6 µA
V
RUN/SS
Run/SS Threshold 0.4 0.7 1.0 V
I
RUN/SS
Soft-Start Current Source V
RUN/SS
= 0V 1.2 2.25 3.3 µA
V
SYNC/FCB
Auxiliary Feedback Threshold V
SYNC/FCB
Ramping Negative 0.730 0.8 0.860 V
I
SYNC/FCB
Auxiliary Feedback Current V
SYNC/FCB
= 0V 0.5 1.5 2.5 µA
f
OSC
Oscillator Frequency V
FB
= 0.8V 315 350 385 kHz
V
FB
= 0V 35 kHz
V
UVLO
Undervoltage Lockout V
IN
Ramping Down from 3V (–40°C to 85°C) 2.3 2.50 2.65 V
V
IN
Ramping Up from 0V (–40°C to 85°C) 2.65 2.85 V
V
IN
Ramping Down from 3V (0°C to 70°C) 2.4 2.50 2.65 V
V
IN
Ramping Up from 0V (0°C to 70°C) 2.65 2.80 V
R
PFET
R
DS(ON)
of P-Channel FET (V
IN
– V
DR
) = 5V, I
SW
= 100mA 0.5 0.7 Ω
R
NFET
R
DS(ON)
of N-Channel FET I
SW
= –100mA 0.6 0.8 Ω
I
LSW
SW Leakage V
RUN/SS
= 0V ±10 ±1000 nA
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: T
J
is calculated from the ambient temperature T
A
and power
dissipation P
D
according to the following formula:
T
J
= T
A
+ (P
D
• 110°C/W)
Note 3: The LTC1627 is tested in a feedback loop that servos V
FB
to the
balance point for the error amplifier (V
ITH
= 0.8V).
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
LTC1627CS8
LTC1627IS8
The ● denotes specifications which apply over the full operating temperature
range, otherwise specifications are at TA = 25°C. VIN = 5V unless otherwise specified.
ELECTRICAL CHARACTERISTICS
ABSOLUTE AXI U RATI GS
WWWU
PACKAGE/ORDER I FOR ATIO
UU
W
3
LTC1627
INPUT VOLTAGE (V)
0
EFFICIENCY (%)
90
95
100
8
1627 G01
85
80
75 24610
V
OUT
= 2.5V
L = 15µH
V
DR
= 0V
Burst Mode OPERATION
I
LOAD
= 100mA
I
LOAD
= 300mA
I
LOAD
= 10mA
Efficiency vs Input Voltage
OUTPUT CURRENT (mA)
1
70
EFFICIENCY (%)
90
95
100
10 100 1000
1627 G03
85
80
75
VIN = 3.6V
VOUT = 2.5V
L = 15µH
VDR = 0V
Burst Mode
OPERATION
SYNCHRONIZED
AT 525kHz
FORCED
CONTINUOUS
OUTPUT CURRENT (mA)
1
70
EFFICIENCY (%)
90
95
100
10 100 1000
1627 G02
85
80
75
V
IN
= 3.6V
V
OUT
= 2.5V
L = 15µH
Burst Mode OPERATION
V
DR
= 0V
V
DR
= –V
IN
Efficiency vs Load Current Efficiency vs Load Current
Undervoltage Lockout Threshold
vs Temperature DC Supply Current*
vs Input VoltageEfficiency vs Load Current
OUTPUT CURRENT (mA)
1
70
EFFICIENCY (%)
90
95
100
10 100 1000
1627 G04
85
80
75
VOUT = 2.5V
L = 15µH
VDR = 0V
Burst Mode OPERATION
VIN = 2.8V
VIN = 3.6V
VIN = 7.2V
TEMPERATURE (°C)
–50 –25
2.30
UNDERVOLTAGE LOCKOUT THRESHOLD (V)
2.35
2.45
2.50
2.55
75 100
2.75
1627 G05
2.40
0 25 50 125
2.60
2.65
2.70
V
IN
RAMPING UP
V
IN
RAMPING DOWN
INPUT VOLTAGE (V)
*DOES NOT INCLUDE GATE CHARGE CURRENT
2.5
100
BATTERY VOLTAGE (V)
150
250
300
350
6.5
550
1627 G06
200
4.5
3.5 7.5
5.5 8.5
400
450
500 T
J
= 25°C
V
OUT
= 1.8V
SYNCHRONIZED AT 525kHz
Burst Mode OPERATION
Reference Voltage
vs Temperature Forced Continuous Threshold
Voltage vs Temperature
Supply Current in Shutdown
vs Input Voltage
TEMPERATURE (°C)
–50 –25
0.790
REFERENCE VOLTAGE (V)
0.792
0.796
0.798
0.800
75 100
0.808
1627 G08
0.794
0 25 50 125
0.802
0.804
0.806 V
IN
= 5V
TEMPERATURE (°C)
–50 –25
0.790
FORCED CONTINUOUS THRESHOLD VOLTAGE (V)
0.792
0.796
0.798
0.800
75 100
0.808
1627 G09
0.794
0 25 50 125
0.802
0.804
0.806 V
IN
= 5V
INPUT VOLTAGE (V)
2.5
4
SUPPLY CURRENT IN SHUTDOWN (µA)
6
10
12
14
6.5
22
1627 G07
8
4.5
3.5 7.5
5.5 8.5
16
18
20 V
RUN/SS
= 0V
T
J
= 85°C
T
J
= 25°C
T
J
= –40°C
TYPICAL PERFOR A CE CHARACTERISTICS
UW
4
LTC1627
Oscillator Frequency
vs Temperature
TEMPERATURE (°C)
–50 –25
300
OSCILLATOR FREQUENCY (kHz)
310
330
340
350
75 100
390
1627 G10
320
0 25 50 125
360
370
380 V
IN
= 5V
V
SYNC/FCB
= 0V
Maximum Output Load Current
vs Input Voltage
INPUT VOLTAGE (V)
2.5
200
MAXIMUM OUTPUT LOAD CURRENT (mA)
300
500
600
700
6.5
1100
1627 G12
400
4.5
3.5 7.5
5.5 8.5
800
900
1000 V
DR
= –V
IN
V
DR
= 0V
V
OUT
= 2.5V
L = 15µH
INPUT VOLTAGE (V)
2.5
300
OSCILLATOR FREQUENCY (kHz)
310
330
340
350
6.5
390
1627 G11
320
4.5
3.5 7.5
5.5 8.5
360
370
380 V
SYNC/FCB
= 0V
Oscillator Frequency
vs Input Voltage
TEMPERATURE (°C)
–50 –25
0
SYNCHRONOUS SWITCH LEAKAGE (nA)
200
600
800
1000
75 100
1800
1627 G13
400
0 25 50 125
1200
1400
1600 V
IN
= 8.4V
V
DR
= 0V
SYNCHRONOUS
SWITCH
MAIN
SWITCH
Switch Leakage Current
vs Temperature Switch Resistance
vs Temperature
TEMPERATURE (°C)
–50 –25
0
SWITCH RESISTANCE (Ω)
0.1
0.3
0.4
0.5
75 100
0.9
1627 G14
0.2
0 25 50 125
0.6
0.7
0.8 V
IN
= 5V
V
DR
= 0V
SYNCHRONOUS
SWITCH
MAIN
SWITCH
Switch Resistance
vs Input Voltage
INPUT VOLTAGE (V)
2.5
0
SWITCH RESISTANCE (Ω)
0.1
0.3
0.4
0.5
6.5
0.9
1627 G15
0.2
4.5
3.5 7.5
5.5 8.5
0.6
0.7
0.8
V
DR
= 0V
SYNCHRONOUS SWITCH
MAIN SWITCH
Load Step Transient Response
V
OUT
50mV/DIV
AC COUPLED
I
TH
0.5V/DIV
I
LOAD
500mA/DIV
25µs/DIV
V
IN
= 5V
V
OUT
= 3.3V
L = 15µH
C
IN
= 22µF
C
OUT
= 100µF
I
LOAD
= 0mA TO 500mA
Burst Mode OPERATION
1627 G16
V
OUT
20mV/DIV
AC COUPLED
I
LOAD
200mA/DIV
10µs/DIV
V
IN
= 5V
V
OUT
= 3.3V
L = 15µH
C
IN
= 22µF
C
OUT
= 100µF
I
LOAD
= 50mA
1627 G18
V
OUT
50mV/DIV
AC COUPLED
I
TH
0.5V/DIV
I
LOAD
500mA/DIV
25µs/DIV
V
IN
= 5V
V
OUT
= 3.3V
L = 15µH
C
IN
= 22µF
C
OUT
= 100µF
I
LOAD
= 0mA TO 500mA
FORCED CONTINUOUS MODE
1627 G17
Load Step Transient Response Burst Mode Operation
SW
5V/DIV
TYPICAL PERFOR A CE CHARACTERISTICS
UW
5
LTC1627
I
TH
(Pin 1): Error Amplifier Compensation Point. The
current comparator threshold increases with this control
voltage. Nominal voltage range for this pin is 0V to 1.2V.
RUN/SS (Pin 2): Combination of Soft-Start and Run
Control Inputs. A capacitor to ground at this pin sets the
ramp time to full current output. The time is approximately
0.5s/µF. Forcing this pin below 0.4V shuts down all the
circuitry.
V
FB
(Pin 3): Feedback Pin. Receives the feedback voltage
from an external resistive divider across the output.
GND (Pin 4): Ground Pin.
SW (Pin 5): Switch Node Connection to Inductor. This pin
connects to the drains of the internal main and synchro-
nous power MOSFET switches.
V
IN
(Pin 6):
Main Supply Pin. Must be closely decoupled
to GND, Pin 4.
V
DR
(Pin 7): Top Driver Return Pin. This pin can be
bootstrapped to go below ground to improve efficiency at
low V
IN
(see Applications Information).
SYNC/FCB (Pin 8): Multifunction Pin. This pin performs
three functions: 1) secondary winding feedback input, 2)
external clock synchronization and 3) Burst Mode opera-
tion or forced continuous mode select. For secondary
winding applications connect a resistive divider from the
secondary output. To synchronize with an external clock
apply a TTL/CMOS compatible clock with a frequency
between 385kHz and 525kHz. To select Burst Mode opera-
tion, float the pin or tie it to V
IN
. Grounding Pin 8 forces
continuous operation (see Applications Information).
V
FB
V
IN
3
2
1
6
7
5
4
Q
Q
R
S
0.12V
SWITCHING
LOGIC
AND
BLANKING
CIRCUIT
0.6V
I
TH
RUN/SS
0.8V
0.4V
0.86V
SYNC/FCB
Y = “0” ONLY WHEN X IS A CONSTANT “1”
SHUTDOWN
SLEEP 6Ω
1.5µA
2.25µA
V
IN
V
IN
V
IN
V
IN
V
DR
SW
GND
1627 BD
V
IN
0.8V
BURST
EN
8OSC
FREQ
SHIFT
0.8V
REF
UVLO
TRIP = 2.5V
Y
X
BURST
DEFEAT
SLOPE
COMP
–
+
OVDET
–
+
–
+
–
+
–+
–
+
–
+
EA I
COMP
–
+
I
RCMP
RUN/SOFT
START
FCB
ANTI-
SHOOT-THRU
UU
U
PI FU CTIO S
FU CTIO AL DIAGRA
UU
W
6
LTC1627
Main Control Loop
The LTC1627 uses a constant frequency, current mode
step-down architecture. Both the main and synchronous
switches, consisting of top P-channel and bottom
N-channel power MOSFETs, are internal. During normal
operation, the internal top power MOSFET is turned on
each cycle when the oscillator sets the RS latch, and
turned off when the current comparator, I
COMP
, resets the
RS latch. The peak inductor current at which I
COMP
resets
the RS latch is controlled by the voltage on the I
TH
pin,
which is the output of error amplifier EA. The V
FB
pin,
described in the Pin Functions section, allows EA to
receive an output feedback voltage from an external resis-
tive divider. When the load current increases, it causes a
slight decrease in the feedback voltage relative to the 0.8V
reference, which, in turn, causes the I
TH
voltage to in-
crease until the average inductor current matches the new
load current. While the top MOSFET is off, the bottom
MOSFET is turned on until either the inductor current
starts to reverse as indicated by the current reversal
comparator I
RCMP
, or the beginning of the next cycle.
The main control loop is shut down by pulling the RUN/SS
pin low. Releasing RUN/SS allows an internal 2.25µA
current source to charge soft-start capacitor C
SS
. When
C
SS
reaches 0.7V, the main control loop is enabled with the
I
TH
voltage clamped at approximately 5% of its maximum
value. As C
SS
continues to charge, I
TH
is gradually
released, allowing normal operation to resume.
Comparator OVDET guards against transient overshoots
>7.5% by turning the main switch off and turning the
synchronous switch on. With the synchronous switch
turned on, the output is crowbarred. This may cause a
large amount of current to flow from V
IN
if the main switch
is damaged, blowing the system fuse.
Burst Mode Operation
The LTC1627 is capable of Burst Mode operation in which
the internal power MOSFETs operate intermittently based
on load demand. To enable Burst Mode operation, simply
allow the SYNC/FCB pin to float or connect it to a logic
high. To disable Burst Mode operation and enable forced
continuous mode, connect the SYNC/FCB pin to GND. In
this mode, the efficiency is lowest at light loads, but
becomes comparable to Burst Mode operation when the
output load exceeds 100mA. The threshold voltage be-
tween Burst Mode operation and forced continuous mode
is 0.8V. This can be used to assist in secondary winding
regulation as described in Auxiliary Winding Control Using
SYNC/FCB Pin in the Applications Information section.
When the converter is in Burst Mode operation, the peak
current of the inductor is set to approximately 200mA,
even though the voltage at the I
TH
pin indicates a lower
value. The voltage at the I
TH
pin drops when the inductor’s
average current is greater than the load requirement. As
the I
TH
voltage drops below 0.12V, the BURST comparator
trips, causing the internal sleep line to go high and turn off
both power MOSFETs.
In sleep mode, both power MOSFETs are held off and the
internal circuitry is partially turned off, reducing the quies-
cent current to 200µA. The load current is now being
supplied from the output capacitor. When the output
voltage drops, causing I
TH
to rise above 0.22V, the top
MOSFET is again turned on and this process repeats.
Short-Circuit Protection
When the output is shorted to ground, the frequency of the
oscillator is reduced to about 35kHz, 1/10 the nominal
frequency. This frequency foldback ensures that the
inductor current has more time to decay, thereby prevent-
ing runaway. The oscillator’s frequency will progressively
increase to 350kHz (or the synchronized frequency) when
V
FB
rises above 0.3V.
Frequency Synchronization
The LTC1627 can be synchronized with an external
TTL/CMOS compatible clock signal. The frequency range
of this signal must be from 385kHz to 525kHz.
Do not
attempt to synchronize the LTC1627 below 385kHz as this
may cause abnormal operation and an undesired fre-
quency spectrum. The top MOSFET turn-on follows the
rising edge of the external source.
When the LTC1627 is clocked by an external source, Burst
Mode operation is disabled; the LTC1627 then operates in
PWM pulse skipping mode. In this mode, when the output
load is very low, current comparator I
COMP
remains tripped
for more than one cycle and forces the main switch to stay
off for the same number of cycles. Increasing the output
(Refer to Functional Diagram)
OPERATIO
U
7
LTC1627
load slightly allows constant frequency PWM operation
to resume.
Frequency synchronization is inhibited when the feedback
voltage V
FB
is below 0.6V. This prevents the external clock
from interfering with the frequency foldback for short-
circuit protection.
Dropout Operation
When the input supply voltage decreases toward the out-
put voltage, the duty cycle increases toward the maximum
on-time. Further reduction of the supply voltage forces the
main switch to remain on for more than one cycle until it
reaches 100% duty cycle. The output voltage will then be
determined by the input voltage minus the voltage drop
across the P-channel MOSFET and the inductor.
In Burst Mode operation or pulse skipping mode operation
(externally synchronized) with the output lightly loaded,
the LTC1627 transitions through continuous mode as it
enters dropout.
Undervoltage Lockout
A precision undervoltage lockout shuts down the LTC1627
when V
IN
drops below 2.5V, making it ideal for single
lithium-ion battery applications. In lockout, the LTC1627
draws only several microamperes, which is low enough to
prevent deep discharge and possible damage to the lithium-
ion battery nearing its end of charge. A 150mV hysteresis
ensures reliable operation with noisy supplies.
Low Supply Operation
The LTC1627 is designed to operate down to 2.65V supply
voltage. At this voltage the converter is most likely to be
running at high duty cycles or in dropout where the main
switch is on continuously. Hence, the I
2
R loss is due
mainly to the R
DS(ON)
of the P-channel MOSFET. See
Efficiency Considerations in the Applications Information
section.
When V
IN
is low (<4.5V) the R
DS(ON)
of the P-channel
MOSFET can be lowered by driving its gate below ground.
The top P-channel MOSFET driver makes use of a floating
return pin, V
DR
, to allow biasing below GND. A simple
charge pump bootstrapped to the SW pin realizes a
negative voltage at the V
DR
pin as shown in Figure 2. Using Figure 3. Maximum Inductor Peak Current vs Duty Cycle
L1
C
OUT
100µF
V
OUT
1627 F02
V
IN
< 4.5V
D1
D2
C1
0.1µF
C2
0.1µF
V
DR
V
IN
SW
LTC1627
+
010 20 30 40 50 60 70 80 90 100
950
900
850
800
750
700
650
600
550
500
DUTY CYCLE (%)
1627 F03
MAXIMUM INDUCTOR PEAK CURRENT (mA)
WORST CASE
EXTERNAL
CLOCK SYNC
WITHOUT
EXTERNAL
CLOCK SYNC
V
IN
= 5V
Figure 2. Using a Charge Pump to Bias VDR
the charge pump at V
IN
≥ 4.5V is not recommended to
ensure that (V
IN
– V
DR
) does not exceed its absolute
maximum voltage.
When V
IN
decreases to a voltage close to V
OUT
, the loop
may enter dropout and attempt to turn on the P-channel
MOSFET continuously. When the V
DR
charge pump is
enabled, a dropout detector counts the number of oscilla-
tor cycles that the P-channel MOSFET remains on, and
periodically forces a brief off period to allow C1 to
recharge. 100% duty cycle is allowed when V
DR
is grounded.
Slope Compensation and Inductor Peak Current
Slope compensation provides stability by preventing
subharmonic oscillations. It works by internally adding a
ramp to the inductor current signal at duty cycles in excess
of 40%. As a result, the maximum inductor peak current
is lower for V
OUT
/V
IN
> 0.4 than when V
OUT
/V
IN
< 0.4. See
the inductor peak current as a function of duty cycle graph
in Figure 3. The worst-case peak current reduction occurs
with the oscillator synchronized at its minimum frequency,
i.e., to a clock just above the oscillator free-running
OPERATIO
U
8
LTC1627
frequency. The actual reduction in average current is less
than for peak current.
The basic LTC1627 application circuit is shown in Figure
1. External component selection is driven by the load
requirement and begins with the selection of L followed by
C
IN
and C
OUT.
Inductor Value Calculation
The inductor selection will depend on the operating fre-
quency of the LTC1627. The internal preset frequency is
350kHz, but can be externally synchronized up to 525kHz.
The operating frequency and inductor selection are inter-
related in that higher operating frequencies allow the use
of smaller inductor and capacitor values. However, oper-
ating at a higher frequency generally results in lower
efficiency because of internal gate charge losses.
The inductor value has a direct effect on ripple current. The
ripple current ∆I
L
decreases with higher inductance or
frequency and increases with higher V
IN
or V
OUT
.
∆IfLVV
V
L OUT OUT
IN
=
()()
−
11
(1)
Accepting larger values of ∆I
L
allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆I
L
= 0.4(I
MAX
).
The inductor value also has an effect on Burst Mode
operation. The transition to low current operation begins
when the inductor current peaks fall to approximately
200mA. Lower inductor values (higher ∆I
L
) will cause this
to occur at lower load currents, which can cause a dip in
efficiency in the upper range of low current operation. In
Burst Mode operation, lower inductance values will cause
the burst frequency to increase.
Inductor Core Selection
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy,
or Kool Mµ
®
cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires
more turns of wire and therefore copper losses will
increase.
Ferrite designs have very low core losses and are preferred
at high switching frequencies, so design goals can con-
centrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that induc-
tance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Kool Mµ (from Magnetics, Inc.) is a very good, low loss
core material for toroids with a “soft” saturation character-
istic. Molypermalloy is slightly more efficient at high
(>200kHz) switching frequencies but quite a bit more
expensive. Toroids are very space efficient, especially
when you can use several layers of wire, while inductors
wound on bobbins are generally easier to surface mount.
New designs for surface mount are available from
Coiltronics, Coilcraft and Sumida.
C
IN
and C
OUT
Selection
In continuous mode, the source current of the top MOSFET
is a square wave of duty cycle V
OUT
/V
IN
. To prevent large
voltage transients, a low ESR input capacitor sized for the
maximum RMS current must be used. The maximum
RMS capacitor current is given by:
CI
VVV
V
IN MAX OUT IN OUT
IN
required I
RMS
≅−
()
[]
12/
This formula has a maximum at V
IN
= 2V
OUT
, where
I
RMS
= I
OUT
/2. This simple worst-case condition is com-
monly used for design because even significant deviations
do not offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on 2000 hours of life.
This makes it advisable to further derate the capacitor, or
choose a capacitor rated at a higher temperature than
required. Several capacitors may also be paralleled to meet
size or height requirements in the design. Always consult the
manufacturer if there is any question.
Kool Mµ is a registered trademark of Magnetics, Inc.
APPLICATIO S I FOR ATIO
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9
LTC1627
Figure 4. Setting the LTC1627 Output Voltage
0.8V ≤ V
OUT
≤ 8.5V
R2
R1
1627 F04
V
FB
GND
LTC1627
Run/Soft-Start Function
The RUN/SS pin is a dual purpose pin that provides the
soft-start function and a means to shut down the LTC1627.
Soft-start reduces surge currents from V
IN
by gradually
increasing the internal current limit. Power supply
sequencing can also be accomplished using this pin.
An internal 2.25µA current source charges up an external
capacitor C
SS
. When the voltage on RUN/SS reaches 0.7V
the LTC1627 begins operating. As the voltage on RUN/SS
continues to ramp from 0.7V to 1.8V, the internal current
limit is also ramped at a proportional linear rate. The
current limit begins at 25mA (at V
RUN/SS
≤ 0.7V) and ends
at the Figure 3 value (V
RUN/SS
≈ 1.8V). The output current
thus ramps up slowly, charging the output capacitor. If
RUN/SS has been pulled all the way to ground, there will
be a delay before the current starts increasing and is given
by:
tC
A
DELAY SS
=07
225
.
.µ
Pulling the RUN/SS pin below 0.4V puts the LTC1627 into
a low quiescent current shutdown (I
Q
< 15µA). This pin can
be driven directly from logic as shown in Figure 5. Diode
D1 in Figure 5 reduces the start delay but allows C
SS
to
ramp up slowly providing the soft-start function. This
diode can be deleted if soft-start is not needed.
RUN/SS
C
SS
D1
3.3V OR 5V
C
SS
RUN/SS
1627 F05
Figure 5. RUN/SS Pin Interfacing
The selection of C
OUT
is driven by the required effective series
resistance (ESR). Typically, once the ESR requirement is
satisfied, the capacitance is adequate for filtering. The output
ripple ∆V
OUT
is determined by:
∆∆V I ESR fC
OUT L OUT
≅+
1
8
where f = operating frequency, C
OUT
= output capacitance
and ∆I
L
= ripple current in the inductor. The output ripple
is highest at maximum input voltage since ∆I
L
increases
with input voltage. For the LTC1627, the general rule for
proper operation is:
C
OUT
required ESR < 0.25Ω
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance through-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR/size
ratio of any aluminum electrolytic at a somewhat higher
price. Once the ESR requirement for C
OUT
has been met,
the RMS current rating generally far exceeds the
I
RIPPLE(P-P)
requirement.
In surface mount applications multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec-
trolytic and dry tantalum capacitors are both available in
surface mount configurations. In the case of tantalum, it is
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS series of surface mount tantalum, available in case
heights ranging from 2mm to 4mm. Other capacitor types
include Sanyo POSCAP, KEMET T510 and T495 series,
Nichicon PL series and Sprague 593D and 595D series.
Consult the manufacturer for other specific recommenda-
tions.
Output Voltage Programming
The output voltage is set by a resistive divider according
to the following formula:
VV
R
R
OUT
=+
08 1 2
1
.
(2)
The external resistive divider is connected to the output,
allowing remote voltage sensing as shown in Figure 4.
APPLICATIO S I FOR ATIO
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10
LTC1627
Auxiliary Winding Control Using SYNC/FCB Pin
The SYNC/FCB pin can be used as a secondary feedback
input to provide a means of regulating a flyback winding
output. When this pin drops below its ground referenced
0.8V threshold, continuous mode operation is forced. In
continuous mode, the main and synchronous MOSFETs
are switched continuously regardless of the load on the
main output.
Synchronous switching removes the normal limitation
that power must be drawn from the inductor primary
winding in order to extract power from auxiliary windings.
With continuous synchronous operation power can be
drawn from the auxiliary windings without regard to the
primary output load.
The secondary output voltage is set by the turns ratio of the
transformer in conjunction with a pair of external resistors
returned to the SYNC/FCB pin as shown in Figure 6. The
secondary regulated voltage V
SEC
in Figure 6 is given by:
VNVV V
R
R
SEC OUT DIODE
≅+
()()
−>+
1081
4
3
.
where N is the turns ratio of the transformer, V
OUT
is the
main output voltage sensed by V
FB
and V
DIODE
is the
voltage drop across the Schottky diode.
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, two main sources usually account for most of the
losses in LTC1627 circuits: V
IN
quiescent current and I
2
R
losses.
1. The V
IN
quiescent current is due to two components:
the DC bias current as given in the electrical character-
istics and the internal main switch and synchronous
switch gate charge currents. The gate charge current
results from switching the gate capacitance of the
internal power MOSFET switches. Each time the gate is
switched from high to low to high again, a packet of
charge dQ moves from V
IN
to ground. The resulting
dQ/dt is the current out of V
IN
that is typically larger
than the DC bias current. In continuous mode, I
GATECHG
= f(Q
T
+ Q
B
) where Q
T
and Q
B
are the gate charges of
the internal top and bottom switches. Both the DC bias
and gate charge losses are proportional to V
IN
and thus
their effects will be more pronounced at higher supply
voltages.
2. I
2
R losses are calculated from the resistances of the
internal switches R
SW
and external inductor R
L
. In
continuous mode the average output current flowing
through inductor L is “chopped” between the main
switch and the synchronous switch. Thus, the series
resistance looking into SW pin from L is a function of
both top and bottom MOSFET R
DS(ON)
and the duty
cycle (DC) as follows:
R
SW
= (R
DS(ON)TOP
)(DC) + (R
DS(ON)BOT
)(1 – DC)
The R
DS(ON)
for both the top and bottom MOSFETs can
be obtained from the Typical Performance Characteris-
tics curves. Thus, to obtain I
2
R losses, simply add R
SW
to R
L
and multiply by the square of the average output
current.
Other losses including C
IN
and C
OUT
ESR dissipative losses,
MOSFET switching losses and inductor core losses generally
account for less than 2% total additional loss.
Figure 6. Secondary Output Loop Connection
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
1µF
1627 F06
R4
R3
C
OUT
V
OUT
V
SEC
L1
1:N
+
+
SYNC/FCB
SW
LTC1627
APPLICATIO S I FOR ATIO
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11
LTC1627
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, V
OUT
immediately shifts by an amount
equal to (∆I
LOAD
• ESR), where ESR is the effective series
resistance of C
OUT
. ∆I
LOAD
also begins to charge or dis-
charge C
OUT
, which generates a feedback error signal. The
regulator loop then acts to return V
OUT
to its steady-state
value. During this recovery time V
OUT
can be monitored for
overshoot or ringing that would indicate a stability prob-
lem. The internal compensation provides adequate com-
pensation for most applications. But if additional compen-
sation is required, the I
TH
pin can be used for external
compensation as shown in Figure 7.
A second, more severe transient is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with C
OUT
, causing a rapid drop in V
OUT
. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive so that
the load rise time is limited to approximately (25 • C
LOAD
).
Thus, a 10µF capacitor would require a 250µs rise time,
limiting the charging current to about 130mA.
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1627. These items are also illustrated graphically in
the layout diagram of Figure 7. Check the following in your
layout:
1. Are the signal and power grounds segregated? The
LTC1627 signal ground consists of the resistive
divider, the optional compensation network (R
C
and
C
C1
), C
SS
and C
C2
. The power ground consists of the
(–) plate of C
IN
, the (–) plate of C
OUT
and Pin 4 of the
LTC1627. The power ground traces should be kept
short, direct and wide. The signal ground and power
ground should converge to a common node in a star-
ground configuration.
2. Does the V
FB
pin connect directly to the feedback
resistors? The resistive divider R1/R2 must be con-
nected between the (+) plate of C
OUT
and signal ground.
Figure 7. LTC1627 Layout Diagram
CC2
CC1
RC
OPTIONAL OPTIONAL
CSS
COUT
CIN
D1
D2
CV
CB
L1
R2
R1
BOLD LINES INDICATE
HIGH CURRENT PATHS
1
2
3
4
8
7
6
5
SYNC/FCB
VDR
VIN
SW
ITH
RUN/SS
VFB
GND
LTC1627
VIN
1627 F07
+
–
VOUT
+
–
+
+
APPLICATIO S I FOR ATIO
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12
LTC1627
3. Does the (+) plate of C
IN
connect to V
IN
as closely as
possible? This capacitor provides the AC current to the
internal power MOSFETs.
4. Keep the switching node SW away from sensitive small-
signal nodes.
Design Example
As a design example, assume the LTC1627 is used in a
single lithium-ion battery-powered cellular phone applica-
tion. The V
IN
will be operating from a maximum of 4.2V
down to about 2.7V. The load current requirement is a
maximum of 0.5A but most of the time it will be on standby
mode, requiring only 2mA. Efficiency at both low and high
load currents is important. Output voltage is 2.5V. With
this information we can calculate L using equation (1),
LfI
VV
V
LOUT OUT
IN
=
()( )
−
11
∆
(3)
Substituting V
OUT
= 2.5V, V
IN
= 4.2V, ∆I
L
= 200mA and
f = 350kHz in equation (3) gives:
LV
kHz mA
V
VH=
()()
−
=
25
350 200 125
42 14 5
..
..µ
A 15µH inductor works well for this application. For good
efficiency choose a 1A inductor with less than 0.25Ω
series resistance.
C
IN
will require an RMS current rating of at least 0.25A at
temperature and C
OUT
will require an ESR of less than
0.25Ω. In most applications, the requirements for these
capacitors are fairly similar.
For the feedback resistors, choose R1 = 80.6k. R2 can then
be calculated from equation (2) to be:
RVRk
OUT
208 1 1 171=−
•=
.; use 169k
Figure 8 shows the complete circuit along with its effi-
ciency curve.
C
SS
0.1µF
C
OUT†
100µF
6.3V
C
IN††
22µF
16V
C1
0.1µF
C2
0.1µF
D2
D1
BAT54S**
15µH*
* SUMIDA CD54-150
** ZETEX BAT54S
†
AVX TPSC107M006R0150
††
AVX TPSC226M016R0375
R2
169k
1%
R1
80.6k
1%
C
ITH
47pF 1
2
3
4
8
7
6
5
SYNC/FCB
V
DR
V
IN
SW
I
TH
RUN/SS
V
FB
GND
LTC1627 V
IN
2.8V TO
4.5V
V
OUT
2.5V
0.5A
1627 F08a
+
+
OUTPUT CURRENT (mA)
EFFICIENCY (%)
1 100 1000
1627 F08b
10
100
95
90
85
80
75
70
65
60
55
50
45
VOUT = 2.5V
VIN = 3.6V
VIN = 4.2V
Figure 8. Single Lithium-Ion to 2.5V/0.5A Regulator
APPLICATIO S I FOR ATIO
WUUU
13
LTC1627
C
ITH
47pF
C
SS
0.1µF
15µH*
1
2
3
4
8
7
6
5
SYNC/FCB
V
DR
V
IN
SW
I
TH
RUN/SS
V
FB
GND
LTC1627
V
IN
= 5V
V
OUT
3.3V
0.5A
1627 TA03
+
+
C
OUT **
100µF
6.3V
C
IN***
22µF
16V
R2
249k
1%
R1
80.6k
1%
* SUMIDA CD54-150
**
AVX TPSC107M006R0150
***
AVX TPSC226M016R0375
5V Input to 3.3V/0.5A Regulator
Double Lithium-Ion to 5V/0.5A Low Dropout Regulator
C
ITH
47pF
C
SS
0.1µF
33µH*
1
2
3
4
8
7
6
5
SYNC/FCB
V
DR
V
IN
SW
I
TH
RUN/SS
V
FB
GND
LTC1627
V
IN
≤ 8.4V
V
OUT
5V
0.5A
1627 TA04
+
+
C
OUT **
100µF
10V
C
IN***
22µF
16V
R2
422k
1%
R1
80.6k
1%
* SUMIDA CD54-330
**
AVX TPSD107M010R0100
***
AVX TPSC226M016R0375
TYPICAL APPLICATIO S
U
14
LTC1627
3.3V Input to 2.5V/0.5A Regulator
CSS
0.1µF
COUT†
100µF
6.3V
CIN††
22µF
16V
C1
0.1µF
C2
0.1µF
D2
D1
BAT54S**
10µH*
* SUMIDA CD54-100
** ZETEX BAT54S
†AVX TPSC107M006R0150
†† AVX TPSC226M016R0375
R2
169k
1%
R1
80.6k
1%
CITH
47pF 1
2
3
4
8
7
6
5
SYNC/FCB
VDR
VIN
SW
ITH
RUN/SS
VFB
GND
LTC1627
VIN = 3.3V
VOUT
2.5V
0.5A
1627 TA05
+
+
Single Lithium-Ion to 1.8V/0.3A Regulator
C
ITH
47pF
C
SS
0.1µF
15µH*
1
2
3
4
8
7
6
5
SYNC/FCB
V
DR
V
IN
SW
I
TH
RUN/SS
V
FB
GND
LTC1627
V
IN
≤ 4.2V
V
OUT
1.8V
0.3A
1627 TA01
+
+
C
OUT **
100µF
6.3V
C
IN***
22µF
16V
R2
100k
1%
R1
80.6k
1%
* SUMIDA CD54-150
**
AVX TPSC107M006R0150
***
AVX TPSC226M016R0375
TYPICAL APPLICATIO S
U
15
LTC1627
Dimensions in inches (millimeters) unless otherwise noted.
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
U
PACKAGE DESCRIPTIO
0.016 – 0.050
(0.406 – 1.270)
0.010 – 0.020
(0.254 – 0.508)× 45°
0°– 8° TYP
0.008 – 0.010
(0.203 – 0.254)
SO8 1298
0.053 – 0.069
(1.346 – 1.752)
0.014 – 0.019
(0.355 – 0.483)
TYP
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
BSC
1234
0.150 – 0.157**
(3.810 – 3.988)
8765
0.189 – 0.197*
(4.801 – 5.004)
0.228 – 0.244
(5.791 – 6.197)
DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
*
**
Double Lithium-Ion to 2.5V/0.5A Regulator
C
ITH
47pF
C
SS
0.1µF
25µH*
1
2
3
4
8
7
6
5
SYNC/FCB
V
DR
V
IN
SW
I
TH
RUN/SS
V
FB
GND
LTC1627
V
IN
≤ 8.4V
V
OUT
2.5V
0.5A
1627 TA01
+
+
C
OUT **
100µF
6.3V
C
IN***
22µF
16V
R2
169k
1%
R1
80.6k
1%
* SUMIDA CD54-250
**
AVX TPSC107M006R0150
***
AVX TPSC226M016R0375
TYPICAL APPLICATIO S
U
16
LTC1627
1627fa LT/TP 0600 2K REV A • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 1998
***22µF
6.3V
D1
MBR0520LT1
D2
††
ZENER
1.8V
* AVX TPSC226M016R0375
** AVX TPSC107M006R0150
*** AVX TAJA226M006R
V
SEC†††
3.3V
100mA
C
ITH
47pF
C
SS
0.1µF
R2
100k
1%
1627 TA02
R1
80.6k
1%
V
IN
≤ 8.5V
C
IN
*
22µF
16V
C
OUT
**
100µF
6.3V
V
OUT
1.8V
0.3A
R4
80.6k
1%
R3
249k
1%
25µH
†
1:1
+
+
+
1
2
3
4
8
7
6
5
SYNC/FCB
V
DR
V
IN
SW
I
TH
RUN/SS
V
FB
GND
LTC1627
†
COILTRONICS CTX25-1
††
MMSZ4678T1
†††
A 10mA MIN LOAD CURRENT
IS RECOMMENDED
Dual Output 1.8V/300mA and 3.3V/100mA Application
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IN
Up to 36V, Fault Protection
LTC1772 Low Input Voltage Current Mode Step-Down DC/DC Controller 550kHz, 6-Pin SOT-23, I
OUT
Up to 5A,
V
IN
from 2.2V to 10V
LTC1877 High Efficiency Monolithic Step-Down Regulator 550kHz, MS8, V
IN
Up to 12V, I
Q
= 10µA, I
OUT
to 600mA
LTC1878 High Efficiency Monolithic Step-Down Regulator 550kHz, MS8, V
IN
Up to 7V, I
Q
= 10µA, I
OUT
to 600mA
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
●
FAX: (408) 434-0507
●
www.linear-tech.com
TYPICAL APPLICATIO S
U
