DP83849IDVSXNOPB - TEXAS INSTRUMENTS

16-Bit 1 MSPS PulSAR®

Unipolar ADC with Reference

AD7667

Rev. 0

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FEATURES

2.5 V internal reference: typical drift 3 ppm/°C

Guaranteed max drift 15 ppm/°C

Throughput:

1 MSPS (Warp mode)

800 kSPS (Normal mode)

666 kSPS (Impulse mode)

INL: ±2.0 LSB max (±0.0038% of full scale)

16-bit resolution with no missing codes

S/(N+D): 88 dB min @ 20 kHz

THD: –96 dB max @ 20 kHz

Analog input voltage range: 0 V to 2.5 V

No pipeline delay

Parallel and serial 5 V/3 V interface

SPI®/QSPITM/MICROWIRETM/DSP compatible

Single 5 V supply operation

Power dissipation

87 mW typ @ 666 kSPS, 130 µW @ 1 kSPS without REF

133 mW typ @1 MSPS with REF

48-lead LQFP and 48-lead LFCSP packages

Pin-to-pin compatible with AD7671, AD7677

APPLICATIONS

Data acquisition

Medical instruments

Digital signal processing

Battery-powered systems

Process control

GENERAL DESCRIPTION

The AD7667 is a 16-bit, 1 MSPS, charge redistribution SAR

analog-to-digital converter that operates from a single 5 V

power supply. The part contains a high speed 16-bit sampling

ADC, an internal conversion clock, internal reference, error

correction circuits, and both serial and parallel system inter-

face ports. It features a very high sampling rate mode (Warp), a

fast mode (Normal) for asynchronous applications, and a

reduced power mode (Impulse) for low power applications

where power is scaled with the throughput. The AD7667 is

hardware factory-calibrated and comprehensively tested to

ensure ac parameters such as signal-to-noise ratio (SNR) and

total harmonic distortion (THD) in addition to the more

traditional dc parameters of gain, offset, and linearity. Operation

is specified from –40°C to +85°C.

FUNCTIONAL BLOCK DIAGRAM

03035-0-001

SWITCHED

CAP DAC

CONTROL LOGIC AND

CALIBRATION CIRCUITRY

CLOCK

AD7667

DATA[15:0]

BUSY

SER/PAR

OB/2C

OGND

OVDD

DGNDDVDD

AVDD

AGND

REF REFGND

INGND

RESET

SERIAL

PORT

PARALLEL

INTERFACE

CNVSTWARP IMPULSE

REF

REFBUFIN

PDBUF

PDREF

BYTESWAP

Figure 1.

Table 1. PulSAR Selection

Type/kSPS 100–250 500–570

800–

1000

Pseudo-

Differential

AD7651

AD7660/AD7661

AD7650/AD7652

AD7664/AD7666

AD7653

AD7667

True Bipolar AD7663 AD7665 AD7671

True

Differential

AD7675 AD7676 AD7677

18-Bit AD7678 AD7679 AD7674

Multichannel/

Simultaneous

AD7654

AD7655

PRODUCT HIGHLIGHTS

1. Fast Throughput.

The AD7667 is a 1 MSPS, charge redistribution, 16-bit SAR

ADC with internal error correction circuitry.

2. Superior INL.

The AD7667 has a maximum integral nonlinearity of

2.0 LSBs with no missing 16-bit codes.

3. Internal Reference.

The AD7667 has an internal reference with a typical

temperature drift of 3 ppm/°C.

4. Single-Supply Operation.

The AD7667 operates from a single 5 V supply. In Impulse

mode, its power dissipation decreases with throughput.

5. Serial or Parallel Interface.

Versatile parallel or 2-wire serial interface arrangement is

compatible with both 3 V and 5 V logic.

AD7667

Rev. 0 | Page 2 of 28

TABLE OF CONTENTS

Specifications..................................................................................... 3

Timing Specifications....................................................................... 5

Absolute Maximum Ratings............................................................ 7

ESD Caution.................................................................................. 7

Pin Configuration and Function Descriptions............................. 8

Definitions of Specifications......................................................... 11

Typical Performance Characteristics ........................................... 12

Circuit Information........................................................................ 16

Converter Operation.................................................................. 16

Typical Connection Diagram ................................................... 18

Power Dissipation versus Throughput .................................... 20

Conversion Control.................................................................... 21

Digital Interface.......................................................................... 22

Parallel Interface......................................................................... 22

Serial Interface............................................................................ 22

Master Serial Interface............................................................... 23

Slave Serial Interface .................................................................. 24

Microprocessor Interfacing....................................................... 26

Application Hints ........................................................................... 27

Bipolar and Wider Input Ranges.............................................. 27

Layout .......................................................................................... 27

Evaluating the AD7667’s Performance.................................... 27

Outline Dimensions....................................................................... 28

Ordering Guide .......................................................................... 28

REVISION HISTORY

Revision 0: Initial Version.

AD7667

Rev. 0 | Page 3 of 28

SPECIFICATIONS

Table 2. –40°C to +85°C, AVDD = DVDD = 5 V, OVDD = 2.7 V to 5.25 V, unless otherwise noted.

Parameter Conditions Min Typ Max Unit

RESOLUTION 16 Bits

ANALOG INPUT

Voltage Range VIN – VINGND 0 VREF V

Operating Input Voltage VIN –0.1 +3 V

INGND –0.1 +0.5 V

Analog Input CMRR fIN = 100 kHz 64 dB

Input Current 1 MSPS Throughput 19 µA

Input Impedance1

THROUGHPUT SPEED

Complete Cycle In Warp Mode 1 µs

Throughput Rate In Warp Mode 1 1000 kSPS

Time between Conversions In Warp Mode 1 ms

Complete Cycle In Normal Mode 1.25 µs

Throughput Rate In Normal Mode 0 800 kSPS

Complete Cycle In Impulse Mode 1.5 µs

Throughput Rate In Impulse Mode 0 666 kSPS

DC ACCURACY

Integral Linearity Error –2.0 +2.0 LSB2

No Missing Codes 16 Bits

Differential Linearity Error –1.0 +1.5 LSB

Transition Noise 0.7 LSB

Unipolar Zero Error, TMIN to TMAX3 ±25 LSB

Unipolar Zero Error Temperature Drift ±1.0 ppm/°C

Full-Scale Error, TMIN to TMAX3 REF = 2.5 V ±0.08 % of FSR

Full-Scale Error Temperature Drift ±1.0 ppm/°C

Power Supply Sensitivity AVDD = 5 V ± 5% ±2 LSB

AC ACCURACY

Signal-to-Noise fIN = 20 kHz 88 89.2 dB4

Spurious Free Dynamic Range fIN = 20 kHz 96 105 dB

Total Harmonic Distortion fIN = 20 kHz –104 –96 dB

Signal-to-(Noise + Distortion) fIN = 20 kHz 88 89 dB

–60 dB Input, fIN = 20 kHz 30 dB

–3 dB Input Bandwidth 13 MHz

SAMPLING DYNAMICS

Aperture Delay 2 ns

Aperture Jitter 5 ps rms

Transient Response Full-Scale Step 250 ns

REFERENCE

Internal Reference Voltage VREF @ 25°C 2.493 2.5 2.507 V

Internal Reference Temperature Drift –40°C to +85°C ±3 ±15 ppm/°C

Output Voltage Hysteresis –40°C to +85°C 50 ppm

Long-Term Drift 100 ppm/1000 Hours

Line Regulation AVDD = 5 V ± 5% ±15 ppm/V

Turn-On Settling Time CREF = 10 µF 5 ms

Temperature Pin

Voltage Output @ 25°C 300 mV

Temperature Sensitivity 1 mV/°C

Output Resistance 4 kΩ

External Reference Voltage Range 2.3 2.5 AVDD – 1.85 V

External Reference Current Drain 1 MSPS Throughput 242 µA

AD7667

Rev. 0 | Page 4 of 28

Parameter Conditions Min Typ Max Unit

DIGITAL INPUTS

Logic Levels

VIL –0.3 +0.8 V

VIH 2.0 DVDD + 0.3 V

IIL –1 +1 µA

IIH –1 +1 µA

DIGITAL OUTPUTS

Data Format5

Pipeline Delay6

VOL I

SINK = 1.6 mA 0.4 V

VOH I

SOURCE = –500 µA OVDD – 0.6 V

POWER SUPPLIES

Specified Performance

AVDD 4.75 5 5.25 V

DVDD 4.75 5 5.25 V

OVDD 2.7 5.257 V

Operating Current8 1 MSPS Throughput

AVDD9 With Reference and Buffer 18.7 mA

AVDD10 Reference and Buffer Alone 3 mA

DVDD11 7.8 mA

OVDD11 200 µA

Power Dissipation without REF9, 11 666 kSPS Throughput 87 115 mW

1 kSPS Throughput 130 µW

Power Dissipation with REF8, 9 1 MSPS Throughput 133 145 mW

TEMPERATURE RANGE12

Specified Performance TMIN to TMAX –40 +85 °C

1See Analog Input section.

2LSB means least significant bit. With the 0 V to 2.5 V input range, 1 LSB is 38.15 µV.

3See Definitions of Specifications section. These specifications do not include the error contribution from the external reference.

4All specifications in dB are referred to a full-scale input FS. Tested with an input signal at 0.5 dB below full-scale, unless otherwise specified.

5Parallel or serial 16-bit.

6Conversion results are available immediately after completed conversion.

7 The max should be the minimum of 5.25 V and DVDD + 0.3 V.

8 In Warp mode.

9 With REF, PDREF and PDBUF are LOW; without REF, PDREF and PDBUF are HIGH.

10 With PDREF, PDBUF LOW and PD HIGH.

11 Impulse mode. Tested in parallel reading mode

12Consult factory for extended temperature range.

AD7667

Rev. 0 | Page 5 of 28

TIMING SPECIFICATIONS

Table 3. –40°C to +85°C, AVDD = DVDD = 5 V, OVDD = 2.7 V to 5.25 V, unless otherwise noted.

Parameter Symbol Min Typ Max Unit

Refer to Figure 33 and Figure 34

Convert Pulse Width t110 ns

Time between Conversions (Warp Mode/Normal Mode/Impulse Mode)1t21/1.25/1.5 µs

CNVST LOW to BUSY HIGH Delay t3 35 ns

BUSY HIGH All Modes except Master Serial Read after Convert t4 0.75/1/1.25 µs

Aperture Delay t5 2 ns

End of Conversion to BUSY LOW Delay t610 ns

Conversion Time t7 0.75/1/1.25 µs

Acquisition Time t8250 ns

RESET Pulse Width t910 ns

Refer to Figure 35, Figure 36, and Figure 37 (Parallel Interface Modes)

CNVST LOW to DATA Valid Delay t10 0.75/1/1.25 µs

DATA Valid to BUSY LOW Delay t11 12 ns

Bus Access Request to DATA Valid t12 45 ns

Bus Relinquish Time t13 5 15 ns

Refer to Figure 39 and Figure 40 (Master Serial Interface Modes)2

CS LOW to SYNC Valid Delay t14 10 ns

CS LOW to Internal SCLK Valid Delay2 t15 10 ns

CS LOW to SDOUT Delay t16 10 ns

CNVST LOW to SYNC Delay t17 25/275/525 ns

SYNC Asserted to SCLK First Edge Delay t18 3 ns

Internal SCLK Period3t19 25 40 ns

Internal SCLK HIGH3 t

20 12 ns

Internal SCLK LOW3 t

21 7 ns

SDOUT Valid Setup Time3 t

22 4 ns

SDOUT Valid Hold Time3 t

23 2 ns

SCLK Last Edge to SYNC Delay3 t

24 3 ns

CS HIGH to SYNC HI-Z t25 10 ns

CS HIGH to Internal SCLK HI-Z t26 10 ns

CS HIGH to SDOUT HI-Z t27 10 ns

BUSY HIGH in Master Serial Read after Convert3 t

28 See Table 4

CNVST LOW to SYNC Asserted Delay t29 0.75/1/1.25 µs

SYNC Deasserted to BUSY LOW Delay t30 25 ns

Refer to Figure 41 and Figure 42 (Slave Serial Interface Modes)2

External SCLK Setup Time t31 5 ns

External SCLK Active Edge to SDOUT Delay t32 3 18 ns

SDIN Setup Time t33 5 ns

SDIN Hold Time t34 5 ns

External SCLK Period t35 25 ns

External SCLK HIGH t36 10 ns

External SCLK LOW t37 10 ns

1In Warp mode only, the time between conversions is 1ms; otherwise there is no required maximum time.

2In serial interface modes, the SYNC, SCLK, and SDOUT timings are defined with a maximum load CL of 10 pF; otherwise, the load is 60 pF maximum.

3In Serial Master Read During Convert mode. See Table 4 for Serial Master Read After Convert Mode.

AD7667

Rev. 0 | Page 6 of 28

Table 4. Serial Clock Timings in Master Read after Convert

DIVSCLK[1] 0 0 1 1

DIVSCLK[0] Symbol 0 1 0 1 Unit

SYNC to SCLK First Edge Delay Minimum t18 3 17 17 17 ns

Internal SCLK Period Minimum t19 25 50 100 200 ns

Internal SCLK Period Maximum t19 40 70 140 280 ns

Internal SCLK HIGH Minimum t20 12 22 50 100 ns

Internal SCLK LOW Minimum t21 7 21 49 99 ns

SDOUT Valid Setup Time Minimum t22 4 18 18 18 ns

SDOUT Valid Hold Time Minimum t23 2 4 30 80 ns

SCLK Last Edge to SYNC Delay Minimum t24 3 55 130 290 ns

BUSY HIGH Width Maximum t24 1.5 2 3 5.25 µs

AD7667

Rev. 0 | Page 7 of 28

ABSOLUTE MAXIMUM RATINGS

Table 5. AD7667 Stress Ratings1

Parameter Rating

IN2, TEMP2, REF, REFBUFIN,

INGND, REFGND to AGND

AVDD + 0.3 V to

AGND – 0.3 V

Ground Voltage Differences

AGND, DGND, OGND ±0.3 V

Supply Voltages

AVDD, DVDD, OVDD –0.3 V to +7 V

AVDD to DVDD, AVDD to OVDD ±7 V

DVDD to OVDD –0.3 V to +7 V

Digital Inputs –0.3 V to DVDD + 0.3 V

PDREF, PDBUF3±20 mA

Internal Power Dissipation4700 mW

Internal Power Dissipation52.5 W

Junction Temperature 150°C

Storage Temperature Range –65°C to +150°C

Lead Temperature Range

(Soldering 10 sec) 300°C

1 Stresses above those listed under Absolute Maximum Ratings may cause

permanent damage to the device. This is a stress rating only; functional

operation of the device at these or any other conditions above those listed

in the operational sections of this specification is not implied. Exposure to

absolute maximum rating conditions for extended periods may affect device

reliability.

2 See section. Analog Input

3 See the Voltage Reference Input section.

4 Specification is for the device in free air:

48-Lead LQFP; θJA = 91°C/W, θJC = 30°C/W.

5 Specification is for the device in free air:

48-Lead LFCSP; θJA = 26°C/W

500µA

1.6mA I

TO OUTPUT

PIN 1.4V

60pF

*IN SERIAL INTERFACE MODES,THE SYNC, SCLK, AND

SDOUT TIMINGS ARE DEFINEDWITH A MAXIMUM LOAD

OF 10pF; OTHERWISE,THE LOAD IS 60pF MAXIMUM.

03033-0-002

Figure 2. Load Circuit for Digital Interface Timing,

SDOUT, SYNC, SCLK Outputs CL = 10 pF

0.8V

DELAY

03033-0-003

Figure 3. Voltage Reference Levels for Timing

ESD CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily

accumulate on the human body and test equipment and can discharge without detection. Although

this product features proprietary ESD protection circuitry, permanent damage may occur on devices

subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are

recommended to avoid performance degradation or loss of functionality.

AD7667

Rev. 0 | Page 8 of 28

PIN CONFIGURATION AND FUNCTION DESCRIPTIONS

13 14 15 16 17 18 19 20 21 22 23 24

48 47 46 45 44 39 38 3743 42 41 40

PIN 1

IDENTIFIER

TOP VIEW

(Not to Scale)

AGND

CNVST

RESET

DGND

AGND

AVDD

BYTESWAP

OB/2C

WARP

IMPULSE

NC = NO CONNECT

SER/PAR

BUSY

D15

D14

D13

AD7667

D3/DIVSCLK1 D12

D4/EXT/INT

D5/INVSYNC

D6/INVSCLK

D7/RDC/SDIN

OGND

OVDD

DVDD

DGND

D8/SDOUT

D9/SCLK

D10/SYNC

D11/RDERROR

PDBUF

PDREF

REFBUFIN

TEMP

AVDD

AGND

INGND

REFGND

REF

03035-0-004

D2/DIVSCLK0

Figure 4. 48-Lead LQFP (ST-48) and 48-Lead LFCSP (CP-48)

Table 6. Pin Function Descriptions

Pin No. Mnemonic Type1Description

1, 36, 41, 42 AGND P Analog Power Ground Pin.

2, 44 AVDD P Input Analog Power Pin. Nominally 5 V.

3, 40 NC No Connect.

6 WARP DI

Mode Selection. When this pin is HIGH and the IMPULSE pin is LOW, this input selects the fastest

mode, the maximum throughput is achievable, and a minimum conversion rate must be applied in

order to guarantee full specified accuracy. When LOW, full accuracy is maintained independent of the

minimum conversion rate.

7 IMPULSE DI

Mode Selection. When IMPULSE is HIGH and WARP is LOW, this input selects a reduced power mode.

In this mode, the power dissipation is approximately proportional to the sampling rate.

4 BYTESWAP DI

Parallel Mode Selection (8-/16-bit). When LOW, the LSB is output on D[7:0] and the MSB is output on

D[15:8]. When HIGH, the LSB is output on D[15:8] and the MSB is output on D[7:0].

5 OB/2C DI Straight Binary/Binary Twos Complement. When OB/2C is HIGH, the digital output is straight binary;

when LOW, the MSB is inverted, resulting in a twos complement output from its internal shift register.

8 SER/PAR DI Serial/Parallel Selection Input. When LOW, the parallel port is selected; when HIGH, the serial interface

mode is selected and some bits of the DATA bus are used as a serial port.

9, 10 D[0:1] DO Bit 0 and Bit 1 of the Parallel Port Data Output Bus. When SER/PAR is HIGH, these outputs are in high

impedance.

11, 12 D[2:3]or

DIVSCLK[0:1]

DI/O When SER/PAR is LOW, these outputs are used as Bit 2 and Bit 3 of the parallel port data output bus.

When SER/PAR is HIGH, EXT/INT is LOW, and RDC/SDIN is LOW (serial master read after convert), these

inputs, part of the serial port, are used to slow down, if desired, the internal serial clock that clocks the

data output. In other serial modes, these pins are not used.

13 D4 or

EXT/INT

DI/O When SER/PAR is LOW, this output is used as Bit 4 of the parallel port data output bus.

When SER/PAR is HIGH, this input, part of the serial port, is used as a digital select input for choosing

the internal data clock or an external data clock. With EXT/INT tied LOW, the internal clock is selected

on the SCLK output. With EXT/INT set to a logic HIGH, output data is synchronized to an external clock

signal connected to the SCLK input.

14 D5 or

INVSYNC

DI/O When SER/PAR is LOW, this output is used as Bit 5 of the parallel port data output bus.

When SER/PAR is HIGH, this input, part of the serial port, is used to select the active state of the SYNC

signal. It is active in both master and slave modes. When LOW, SYNC is active HIGH. When HIGH, SYNC

is active LOW.

15 D6 or

INVSCLK

DI/O When SER/PAR is LOW, this output is used as Bit 6 of the parallel port data output bus.

When SER/PAR is HIGH, this input, part of the serial port, is used to invert the SCLK signal. It is active in

both master and slave modes.

AD7667

Rev. 0 | Page 9 of 28

Pin No. Mnemonic Type1Description

16 D7 or

RDC/SDIN

DI/O When SER/PAR is LOW, this output is used as Bit 7 of the parallel port data output bus.

When SER/PAR is HIGH, this input, part of the serial port, is used as either an external data input or a

read mode selection input depending on the state of EXT/INT.

When EXT/INT is HIGH, RDC/SDIN could be used as a data input to daisy-chain the conversion results

from two or more ADCs onto a single SDOUT line. The digital data level on SDIN is output on DATA

with a delay of 16 SCLK periods after the initiation of the read sequence.

When EXT/INT is LOW, RDC/SDIN is used to select the read mode. When RDC/SDIN is HIGH, the data is

output on SDOUT during conversion. When RDC/SDIN is LOW, the data can be output on SDOUT only

when the conversion is complete.

17 OGND P Input/Output Interface Digital Power Ground.

18 OVDD P Input/Output Interface Digital Power. Nominally at the same supply as the host interface (5 V or 3 V).

19 DVDD P Digital Power. Nominally at 5 V.

20 DGND P Digital Power Ground.

21 D8 or

SDOUT

DO When SER/PAR is LOW, this output is used as Bit 8 of the parallel port data output bus.

When SER/PAR is HIGH, this output, part of the serial port, is used as a serial data output synchronized

to SCLK. Conversion results are stored in an on-chip register. The AD7667 provides the conversion

result, MSB first, from its internal shift register. The DATA format is determined by the logic level of

OB/2C. In serial mode when EXT/INT is LOW, SDOUT is valid on both edges of SCLK. In serial mode

when EXT/INT is HIGH, if INVSCLK is LOW, SDOUT is updated on the SCLK rising edge and valid on the

next falling edge; if INVSCLK is HIGH, SDOUT is updated on the SCLK falling edge and valid on the next

rising edge.

22 D9 or

SCLK

DI/O When SER/PAR is LOW, this output is used as Bit 9 of the parallel port data or SCLK output bus.

When SER/PAR is HIGH, this pin, part of the serial port, is used as a serial data clock input or output,

depending upon the logic state of the EXT/INT pin. The active edge where the data SDOUT is updated

depends upon the logic state of the INVSCLK pin.

23 D10 or

SYNC

DO When SER/PAR is LOW, this output is used as Bit 10 of the parallel port data output bus.

When SER/PAR is HIGH, this output, part of the serial port, is used as a digital output frame

synchronization for use with the internal data clock (EXT/INT = logic LOW). When a read sequence is

initiated and INVSYNC is LOW, SYNC is driven HIGH and remains HIGH while the SDOUT output is

valid. When a read sequence is initiated and INVSYNC is HIGH, SYNC is driven LOW and remains LOW

while the SDOUT output is valid.

24 D11 or

RDERROR

DO When SER/PAR is LOW, this output is used as Bit 11 of the parallel port data output bus. When

SER/PAR and EXT/INT are HIGH, this output, part of the serial port, is used as an incomplete read error

flag. In slave mode, when a data read is started and not complete when the following conversion is

complete, the current data is lost and RDERROR is pulsed HIGH.

25–28 D[12:15] DO

Bit 12 to Bit 15 of the Parallel Port Data Output Bus. These pins are always outputs regardless of the

state of SER/PAR.

29 BUSY DO

Busy Output. Transitions HIGH when a conversion is started and remains HIGH until the conversion is

complete and the data is latched into the on-chip shift register. The falling edge of BUSY could be

used as a data ready clock signal.

30 DGND P Must Be Tied to Digital Ground.

31 RD DI Read Data. When CS and RD are both LOW, the interface parallel or serial output bus is enabled.

32 CS DI Chip Select. When CS and RD are both LOW, the interface parallel or serial output bus is enabled. CS is

also used to gate the external clock.

33 RESET DI

Reset Input. When set to a logic HIGH, this pin resets the AD7667 and the current conversion, if any, is

aborted. If not used, this pin could be tied to DGND.

34 PD DI

Power-Down Input. When set to a logic HIGH, power consumption is reduced and conversions are

inhibited after the current one is completed.

35 CNVST DI Start Conversion. If CNVST is HIGH when the acquisition phase (t8) is complete, the next falling edge

on CNVST puts the internal sample/hold into the hold state and initiates a conversion. The mode is

most appropriate if low sampling jitter is desired. If CNVST is LOW when the acquisition phase (t8) is

complete, the internal sample/hold is put into the hold state and a conversion is immediately started.

37 REF AI/O Reference Input Voltage. On-chip reference output voltage.

38 REFGND AI Reference Input Analog Ground.

39 INGND AI Analog Input Ground.

43 IN AI Primary Analog Input with a Range of 0 V to 2.5 V.

AD7667

Rev. 0 | Page 10 of 28

Pin No. Mnemonic Type1Description

45 TEMP AO Temperature Sensor Voltage Output.

46 REFBUFIN AI/O Reference Input Voltage. The reference output and the reference buffer input.

47 PDREF DI

This pin allows the choice of internal or external voltage references. When LOW, the on-chip reference

is turned on. When HIGH, the internal reference is switched off and an external reference must be

used.

48 PDBUF DI

This pin allows the choice of buffering an internal or external reference with the internal buffer. When

LOW, the buffer is selected. When HIGH, the buffer is switched off.

1AI = Analog Input; AI/O = Bidirectional Analog; AO = Analog Output; DI = Digital Input; DI/O = Bidirectional Digital; DO = Digital Output; P = Power.

AD7667

Rev. 0 | Page 11 of 28

DEFINITIONS OF SPECIFICATIONS

Integral Nonlinearity Error (INL)

Linearity error refers to the deviation of each individual code

from a line drawn from negative full scale through positive full

scale. The point used as negative full scale occurs ½ LSB before

the first code transition. Positive full scale is defined as a level

1½ LSB beyond the last code transition. The deviation is

measured from the middle of each code to the true straight line.

Differential Nonlinearity Error (DNL)

In an ideal ADC, code transitions are 1 LSB apart. Differential

nonlinearity is the maximum deviation from this ideal value. It

is often specified in terms of resolution for which no missing

codes are guaranteed.

Full-Scale Error

The last transition (from 011…10 to 011…11 in twos comple-

ment coding) should occur for an analog voltage 1½ LSB below

the nominal full scale (2.49994278 V for the 0 V to 2.5 V range).

The full-scale error is the deviation of the actual level of the last

transition from the ideal level.

Unipolar Zero Error

The first transition should occur at a level ½ LSB above analog

ground (19.073 µV for the 0 V to 2.5 V range). Unipolar zero

error is the deviation of the actual transition from that point.

Spurious-Free Dynamic Range (SFDR)

SFDR is the difference, in decibels (dB), between the rms

amplitude of the input signal and the peak spurious signal.

Effective Number of Bits (ENOB)

ENOB is a measurement of the resolution with a sine wave

input. It is related to S/(N+D) and is expressed in bits by the

following formula:

ENOB = (S/[N+D]dB – 1.76)/6.02

Total Harmonic Distortion (THD)

THD is the ratio of the rms sum of the first five harmonic

components to the rms value of a full-scale input signal, and is

expressed in decibels.

Signal-to-Noise Ratio (SNR)

SNR is the ratio of the rms value of the actual input signal to the

rms sum of all other spectral components below the Nyquist

frequency, excluding harmonics and dc. The value for SNR is

expressed in decibels.

Signal-to-(Noise + Distortion) Ratio (S/[N+D])

S/(N+D) is the ratio of the rms value of the actual input signal

to the rms sum of all other spectral components below the

Nyquist frequency, including harmonics but excluding dc. The

value for S/(N+D) is expressed in decibels.

Aperture Delay

Aperture delay is a measure of the acquisition performance and

is measured from the falling edge of the CNVST input to when

the input signal is held for a conversion.

Transient Response

Transient response is the time required for the AD7667 to

achieve its rated accuracy after a full-scale step function is

applied to its input.

Reference Voltage Temperature Coefficient

Reference voltage temperature coefficient is derived from the

maximum and minimum reference output voltage (VREF)

measured at TMIN, T(25°C), and TMAX. It is expressed in ppm/°C

using the following equation:

)–()C25(

)–)

)Cppm/( ×

×°

=°

MIN

MAX

REF

REFREF

REF TTV

Min(VMax(V

TCV

where:

VREF(Max) = Maximum VREF at TMIN, T(25°C), or TMAX

VREF(Min) = Minimum VREF at TMIN, T(25°C), or TMAX

VREF(25°C) = VREF at 25°C

TMAX = +85°C

TMIN = –40°C

Thermal Hysteresis

Thermal hysteresis is defined as the absolute maximum change

of reference output voltage after the device is cycled through

temperature from either

T_HYS+ = 25°C to TMAX to 25°C

T_HYS– = 25°C to TMIN to 25°C

It is expressed in ppm using the following equation:

)25(

)_()25(

)( ×

−°

=CV

HYSTVCV

ppmV

REF

REFREF

HYS

where:

VREF(25°C) = VREF at 25°C

VREF(T_HYS) = Maximum change of VREF at T_HYS+ or

T_HYS–

AD7667

Rev. 0 | Page 12 of 28

TYPICAL PERFORMANCE CHARACTERISTICS

03035-0-005

–2.0

–1.5

–1.0

–0.5

0.5

1.0

1.5

2.0

INL (LSB)

0 16384 32768 49152 65536

CODE

Figure 5. Integral Nonlinearity vs. Code

03035-0-006

0 0.5 1.0 1.5 2.0

POSITIVE INL (LSB)

NUMBER OF UNITS

Figure 6. Typical Positive INL Distribution (102 Units)

03035-0-007

NUMBER OF UNITS

0 0.25 0.50 0.75 1.00 1.25 1.50

POSITIVE DNL (LSB)

Figure 7. Typical Positive DNL Distribution (102 Units)

03035-0-008

–1.0

–0.5

0.5

1.0

1.5

0 16384 32768 49152 65536

CODE

DNL (LSB)

Figure 8. Differential Nonlinearity vs. Code

03035-0-009

–2.0 –1.5 –1.0 –0.5 0

NEGATIVE INL (LSB)

NUMBER OF UNITS

Figure 9. Typical Negative INL Distribution (102 Units)

03035-0-010

–1.00 –0.75 –0.50 –0.25 0

NEGATIVE DNL (LSB)

NUMBER OF UNITS

Figure 10. Typical Negative DNL Distribution (102 Units)

AD7667

Rev. 0 | Page 13 of 28

03035-0-011

1344

14654

308

113049

15970

116794

20000

40000

60000

80000

100000

120000

140000

COUNTS

7FFE 7FFF 8000 8001 8002 8003 8004 8005

CODE IN HEX

Figure 11. Histogram of 261,120 Conversions of a

DC Input at the Code Transition

03035-0-012

–180

–160

–140

–120

–100

–80

–60

–40

–20

AMPLITUDE (dB of Full Scale)

0 100 200 300 400 500

FREQUENCY (kHz)

= 1MSPS

= 101.11kHz

SNR = 88.9dB

THD = –94.2dB

SFDR = 94.7dB

S/[N+D] = 88dB

Figure 12. FFT Plot

03035-0-013

SNR

S/[N+D]

ENOB

SNR, S/[N+D] (dB)

1 10 100 1000

FREQUENCY (kHz)

13.0

13.5

14.0

14.5

15.0

15.5

ENOB (Bits)

Figure 13. SNR, S/(N+D), and ENOB vs. Frequency

03035-0-014

035 2602 2958 16 0

51940

56646

146923

20000

40000

60000

80000

100000

120000

140000

160000

180000

COUNTS

7FFE 7FFF 8000 8001 8002 8003 8004 8005 8006

CODE IN HEX

Figure 14. Histogram of 261,120 Conversions of a

DC Input at the Code Center

03035-0-015

SFDR

THD

SECOND

HARMONIC

THIRD

HARMONIC

–120

–115

–110

–105

–100

–95

–90

–85

–80

–75

–70

THD, HARMONICS (dB)

1 10 100 1000

FREQUENCY (kHz)

100

110

120

SFDR (dB)

Figure 15. THD, Harmonics, and SFDR vs. Frequency

03035-0-016

–60 –50 –40 –30 –20 –10 0

INPUT LEVEL (dB)

SNR, S/[N+D] REFERRED TO FULL-SCALE (dB)

SNR

S/[N+D]

Figure 16. SNR and S/[N+D] vs. Input Level (Referred to Full Scale)

AD7667

Rev. 0 | Page 14 of 28

03035-0-017

12.5

13.0

13.5

14.0

14.5

15.0

ENOB (Bits)

SNR, S/[N+D] (dB)

–55 –35 –15 5 25 45 65 85 105 125

SNR

S/[N+D]

ENOB

TEMPERATURE (°C)

Figure 17. SNR, S/[N+D], and ENOB vs. Temperature

03035-0-018

–115

–110

–105

–100

–55 –35 –15 5 25 45 65 85 105 125

TEMPERATURE (°C)

THD, HARMONICS (dB)

THD

SECOND

HARMONIC

THIRD

HARMONIC

Figure 18. THD and Harmonics vs. Temperature

03035-0-019

0.001

0.01

0.1

100

1000

10000

100000

10 100 1k 10k 100k 1M

SAMPLING RATE (SPS)

OPERATING CURRENTS (µA)

AVDD, WARP/NORMAL

DVDD, IMPULSE OVDD, ALL MODES

DVDD, WARP/NORMAL

AVDD, IMPULSE

PDREF = PDBUF = HIGH

Figure 19. Operating Current vs. Sample Rate

03035-0-020

FULL-SCALE

ERROR

ZERO ERROR

–6

–5

–4

–3

–2

–1

ZERO ERROR, FULL-SCALE ERROR (LSB)

–55 –35 –15 5 25 45 65 85 105 125

TEMPERATURE (°C)

Figure 20. Zero Error, Full-Scale Error with Reference vs. Temperature

03035-0-021

2.4966

2.4968

2.4970

2.4972

2.4974

2.4976

2.4978

2.4980

2.4982

2.4984

2.4986

2.4988

–40 –20 0 20 40 60 80 100 120

TEMPERATURE (°C)

VREF (V)

Figure 21. Typical Reference Voltage Output vs. Temperature (3 Units)

03035-0-022

NUMBER OF UNITS

0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0

REFERENCE DRIFT (ppm/°C)

Figure 22. Reference Voltage Temperature Coefficient Distribution (102 Units)

AD7667

Rev. 0 | Page 15 of 28

03035-0-023

0 50 100 150 200

(pF)

DELAY (ns)

OVDD = 2.7V @ 25°C

OVDD = 2.7V @ 85°C

OVDD = 5V @ 85°C

OVDD = 5V @ 25°C

Figure 23. Typical Delay vs. Load Capacitance CL

AD7667

Rev. 0 | Page 16 of 28

CIRCUIT INFORMATION

COMP

REF

REFGND

LSB

MSB

32,768C

INGND

16,384C 4C 2C C C

65,536C

CONTROL

LOGIC

SWITCHES

CONTROL

BUSY

OUTPUT

CODE

03033-0-020

CNVST

Figure 24. ADC Simplified Schematic

The AD7667 is a very fast, low power, single supply, precise

16-bit analog-to-digital converter (ADC). The AD7667 features

different modes to optimize performance according to the

applications. In Warp mode, the part can convert 1 million

samples per second (1 MSPS).

The AD7667 provides the user with an on-chip track/hold,

successive approximation ADC that does not exhibit any

pipeline or latency, making it ideal for multiple multiplexed

channel applications.

The AD7667 can be operated from a single 5 V supply and can

be interfaced to either 5 V or 3 V digital logic. It is housed in

either a 48-lead LQFP or a 48-lead LFCSP that saves space and

allows flexible configurations as either a serial or parallel inter-

face. The AD7667 is pin-to-pin compatible with PulSAR ADCs

and is an upgrade of the AD7666 and AD7661.

CONVERTER OPERATION

The AD7667 is a successive-approximation ADC based on a

charge redistribution DAC. Figure 24 shows a simplified sche-

matic of the ADC. The capacitive DAC consists of an array of

16 binary weighted capacitors and an additional LSB capacitor.

The comparator’s negative input is connected to a dummy

capacitor of the same value as the capacitive DAC array.

During the acquisition phase, the common terminal of the array

tied to the comparator’s positive input is connected to AGND

via SWA. All independent switches are connected to the analog

input IN. Thus, the capacitor array is used as a sampling

capacitor and acquires the analog signal on IN. Similarly, the

dummy capacitor acquires the analog signal on INGND.

When CNVST goes LOW, a conversion phase is initiated. When

the conversion phase begins, SWA and SWB are opened. The

capacitor array and dummy capacitor are then disconnected

from the inputs and connected to REFGND. Therefore, the

differential voltage between IN and INGND captured at the end

of the acquisition phase is applied to the comparator inputs,

causing the comparator to become unbalanced. By switching

each element of the capacitor array between REFGND and REF,

the comparator input varies by binary weighted voltage steps

(VREF/2, VREF/4, …VREF/65536). The control logic toggles these

switches, starting with the MSB, to bring the comparator back

into a balanced condition.

After this process is completed, the control logic generates the

ADC output code and brings the BUSY output LOW.

Modes Of Operation

The AD7667 features three modes of operation: Warp, Normal,

and Impulse. Each mode is best suited for specific applications.

Warp mode allows the fastest conversion rate, up to 1 MSPS.

However in this mode and this mode only, the full specified

accuracy is guaranteed only when the time between conversions

does not exceed 1 ms. If the time between two consecutive

conversions is longer than 1 ms (e.g., after power-up), the first

conversion result should be ignored. This mode makes the

AD7667 ideal for applications where both high accuracy and

fast sample rate are required.

Normal mode is the fastest mode (800 kSPS) without any

limitations on the time between conversions. This mode makes

the AD7667 ideal for asynchronous applications such as data

acquisition systems, where both high accuracy and fast sample

rate are required.

Impulse mode, the lowest power dissipation mode, allows power

saving between conversions. When operating at 1 kSPS, for

example, it typically consumes only 130 µW. This feature makes

the AD7667 ideal for battery-powered applications.

AD7667

Rev. 0 | Page 17 of 28

Transfer Functions Table 7. Output Codes and Ideal Input Voltages

Digital Output Code (Hex)

Description

Analog

Input

Straight

Binary

Twos

Complement

FSR –1 LSB 2.499962 V FFFF17FFF1

FSR – 2 LSB 2.499923 V FFFE 7FFE

Midscale + 1 LSB 1.250038 V 8001 0001

Midscale 1.25 V 8000 0000

Midscale – 1 LSB 1.249962 V 7FFF FFFF

–FSR + 1 LSB 38 µV 0001 8001

–FSR 0 V 00002 80002

Using the OB/2C digital input, the AD7667 offers two output

codings: straight binary and twos complement. The LSB size is

VREF/65536, which is about 38.15 µV. The AD7667’s ideal

transfer characteristic is shown in Figure 25 and Table 7.

000...000

000...001

000...010

111...101

111...110

111...111

ADC CODE (Straight Binary)

ANALOG INPUT

REF

– 1.5 LSB

REF

– 1 LSB

1LSB0V

0.5 LSB

1 LSB =V

REF

/65536

03033-0-021

1This is also the code for overrange analog input (VIN – VINGND above

VREF – VREFGND).

2This is also the code for underrange analog input (VIN below VINGND).

Figure 25. ADC Ideal Transfer Function

NOTES

THE CONFIGURATION SHOWN IS USING THE INTERNAL REFERENCE AND INTERNAL BUFFER.

THE AD8021 IS RECOMMENDED. SEE DRIVER AMPLIFIER CHOICE SECTION.

OPTIONAL LOW JITTER.

A 10

F CERAMIC CAPACITOR (X5R, 1206 SIZE) IS RECOMMENDED (e.g., PANASONIC ECJ3YB0J106M).

SEE VOLTAGE REFERENCE INPUT SECTION.

AD7667

CLOCK

µC/µP/DSP

SERIAL

PORT

DIGITAL SUPPLY

(3.3V OR 5V)

DVDD

100nF +10

100nF

+10

Ω

100nF

+10

ANALOG

SUPPLY

(5V)

2.7nF

15Ω

ANALOG INPUT

(0VTO 2.5V)

PD RESET

IMPULSE

WARP

SER/PAR

OB/2C

BUSY

SDOUT

SCLK

INGND

REFGND

REF

GND

AVDD DGND DVDD OVDD OGND

03035-0-026

PDREF PDBUF RD

BYTESWAP

CNVST

REFBUFIN

100nF

Figure 26. Typical Connection Diagram

AD7667

Rev. 0 | Page 18 of 28

TYPICAL CONNECTION DIAGRAM

Figure 26 shows a typical connection diagram for the AD7667.

Analog Input

Figure 27 shows an equivalent circuit of the input structure of

the AD7667.

The two diodes, D1 and D2, provide ESD protection for the

analog inputs IN and INGND. Care must be taken to ensure

that the analog input signal never exceeds the supply rails by

more than 0.3 V. This will cause these diodes to become

forward-biased and start conducting current. These diodes can

handle a forward-biased current of 100 mA maximum. For

instance, these conditions could eventually occur when the

input buffer’s (U1) supplies are different from AVDD. In such a

case, an input buffer with a short-circuit current limitation can

be used to protect the part.

OR INGND

AGND

AVDD

03033-0-023

Figure 27. Equivalent Analog Input Circuit

This analog input structure allows the sampling of the differen-

tial signal between IN and INGND. Unlike other converters,

INGND is sampled at the same time as IN. By using this

differential input, small signals common to both inputs are

rejected, as shown in Figure 28, which represents the typical

CMRR over frequency with on-chip and external references.

For instance, by using INGND to sense a remote signal ground,

ground potential differences between the sensor and the local

ADC ground are eliminated.

03035-0-028

1 10 100 1000 10000

FREQUENCY (kHz)

CMRR (dB)

EXT REF

REF

Figure 28. Analog Input CMRR vs. Frequency

During the acquisition phase, the impedance of the analog

input IN can be modeled as a parallel combination of capacitor

C1 and the network formed by the series connection of R1 and

C2. C1 is primarily the pin capacitance. R1 is typically 168 Ω

and is a lumped component made up of some serial resistors

and the on resistance of the switches. C2 is typically 60 pF and

is mainly the ADC sampling capacitor. During the conversion

phase, where the switches are opened, the input impedance is

limited to C1. R1 and C2 make a 1-pole low-pass filter that

reduces undesirable aliasing effect and limits the noise.

When the source impedance of the driving circuit is low, the

AD7667 can be driven directly. Large source impedances

signifcantly affect the ac performance, especially total harmonic

distortion (THD). The maximum source impedance depends

on the amount of THD that can be tolerated. The THD

degrades as a function of the source impedance and the

maximum input frequency, as shown in Figure 29.

03035-0-029

–110

–100

–90

–80

–70

–60

–50

THD (dB)

1 10 100 1000

INPUT FREQUENCY (kHz)

= 20

Ω

= 500

Ω

= 50

Ω

= 100

Ω

Figure 29. THD vs. Analog Input Frequency and Source Resistance

Driver Amplifier Choice

Although the AD7667 is easy to drive, the driver amplifier

needs to meet the following requirements:

• The driver amplifier and the AD7667 analog input circuit

must be able to settle for a full-scale step of the capacitor

array at a 16-bit level (0.0015%). In the amplifier’s data

sheet, settling at 0.1% to 0.01% is more commonly speci-

fied. This could differ significantly from the settling time at

a 16-bit level and should be verified prior to driver

selection. The tiny op amp AD8021, which combines ultra

low noise and high gain-bandwidth, meets this settling

time requirement even when used with gains up to 13.

AD7667

Rev. 0 | Page 19 of 28

• The noise generated by the driver amplifier needs to be

kept as low as possible in order to preserve the SNR and

transition noise performance of the AD7667. The noise

coming from the driver is filtered by the AD7667 analog

input circuit 1-pole low-pass filter made by R1 and C2 or

by the external filter, if one is used. The SNR degradation

due to the amplifier is

⎟

⎠

⎞

⎜

⎝

⎛

−2

3)(

784

log20

LOSS

Nef

SNR

where:

f–3dB is the input bandwidth of the AD7667 (13 MHz) or

the cutoff frequency of the input filter (3.9 MHz), if

one is used.

N is the noise factor of the amplifier (+1 in buffer

configuration).

eN is the equivalent input noise voltage of the op amp, in

nV/√Hz.

For instance, a driver with an equivalent input noise of

2 nV/√Hz, like the AD8021 with a noise gain of +1 when

configured as a buffer, degrades the SNR by only 0.13 dB

when using the filter in Figure 26 and by 0.43 dB without.

• The driver needs to have a THD performance suitable to

that of the AD7667. Figure 15 gives the THD versus

frequency that the driver should exceed.

The AD8021 meets these requirements and is appropriate for

almost all applications. The AD8021 needs a 10 pF external

compensation capacitor that should have good linearity as an

NPO ceramic or mica type. Moreover, the use of a noninverting

+1 gain arrangement is recommended and helps to obtain the

best signal-to-noise ratio.

The AD8022 could also be used if a dual version is needed and

gain of 1 is present. The AD829 is an alternative in applications

where high frequency (above 100 kHz) performance is not

required. In gain of 1 applications, it requires an 82 pF

compensation capacitor. The AD8610 is an option when low

bias current is needed in low frequency applications.

Voltage Reference Input

The AD7667 allows the choice of either a very low temperature

drift internal voltage reference or an external 2.5 V reference.

Unlike many ADCs with internal references, the internal

reference of the AD7667 provides excellent performance and

can be used in almost all applications.

To use the internal reference along with the internal buffer,

PDREF and PDBUF should both be LOW. This produces 1.2 V

on REFBUFIN which, amplified by the buffer, results in a 2.5 V

reference on the REF pin.

The output impedance of REFBUFIN is 11 kΩ (minimum)

when the internal reference is enabled. It is necessary to

decouple REFBUFIN with a ceramic capacitor greater than

10 nF. Thus the capacitor provides an RC filter for noise

reduction.

To use an external reference along with the internal buffer,

PDREF should be HIGH and PDBUF should be LOW. This

powers down the internal reference and allows the 2.5 V

reference to be applied to REFBUFIN.

To use an external reference directly on REF pin, PDREF and

PDBUF should both be HIGH.

PDREF and PDBUF power down the internal reference and the

internal reference buffer, respectively. Note that the PDREF and

PDBUF input current should never exceed 20 mA. This could

eventually occur when input voltage is above AVDD (for

instance, at power-up). In this case, a 100 Ω series resistor is

recommended.

The internal reference is temperature compensated to 2.5 V

±7 mV. The reference is trimmed to provide a typical drift of

3 ppm/°C . This typical drift characteristic is shown in

Figure 22. For improved drift performance, an external

reference, such as the AD780, can be used.

The AD7667 voltage reference input REF has a dynamic input

impedance; it should therefore be driven by a low impedance

source with efficient decoupling between the REF and

REFGND inputs. This decoupling depends on the choice of the

voltage reference but usually consists of a low ESR tantalum

capacitor connected to REF and REFGND with minimum

parasitic inductance. A 10 µF (X5R, 1206 size) ceramic chip

capacitor (or 47 µF tantalum capacitor) is appropriate when

using either the internal reference or one of these

recommended reference voltages:

• The low noise, low temperature drift ADR421 and AD780

• The low power ADR291

• The low cost AD1582

AD7667

Rev. 0 | Page 20 of 28

For applications that use multiple AD7667s, it is more effective

to use the internal buffer to buffer the reference voltage.

Care should be taken with the voltage reference’s temperature

coefficient, which directly affects the full-scale accuracy if this

parameter matters. For instance, a ±15 ppm/°C temperature

coefficient of the reference changes full scale by ±1 LSB/°C.

Note that VREF can be increased to AVDD – 1.85 V. Since the

input range is defined in terms of VREF, this would essentially

increase the range to 0 V to 3 V with an AVDD above 4.85 V.

The AD780 can be selected with a 3 V reference voltage.

The TEMP pin, which measures the temperature of the

AD7667, can be used as shown in Figure 30. The output of

TEMP pin is applied to one of the inputs of the analog switch

(e.g., ADG779), and the ADC itself is used to measure its own

temperature. This configuration is very useful for improving the

calibration accuracy over the temperature range.

ADG779

AD8021 C

03035-0-024

ANALOG INPUT

(UNIPOLAR)

AD7667

IN TEMPERATURE

SENSOR

TEMP

Figure 30. Temperature Sensor Connection Diagram

Power Supply

The AD7667 uses three power supply pins: an analog 5 V supply

AVDD, a digital 5 V core supply DVDD, and a digital input/

output interface supply OVDD. OVDD allows direct interface

with any logic between 2.7 V and DVDD + 0.3 V. To reduce the

supplies needed, the digital core (DVDD) can be supplied

through a simple RC filter from the analog supply, as shown in

Figure 26. The AD7667 is independent of power supply

sequencing once OVDD does not exceed DVDD by more than

0.3 V, and is thus free of supply voltage induced latch-up.

Additionally, it is very insensitive to power supply variations

over a wide frequency range, as shown in Figure 31, which

represents PSRR over frequency with on chip and external

references.

03035-0-031

INT REF

EXT REF

1 10 100 1000 10000

FREQUENCY (kHz)

PSRR (dB)

Figure 31. PSRR vs. Frequency

POWER DISSIPATION VERSUS THROUGHPUT

When using the Impulse mode of operation (IMPULSE =

HIGH, WARP = LOW), operating currents are very low during

the acquisition phase, allowing significant power savings when

the conversion rate is reduced (see Figure 32). The AD7667

automatically reduces its power consumption at the end of each

conversion phase. This makes the part ideal for very low power

battery applications. The digital interface and the reference

remain active even during the acquisition phase. To reduce

operating digital supply currents even further, digital inputs

need to be driven close to the power supply rails (i.e., DVDD or

DGND), and OVDD should not exceed DVDD by more than

0.3 V.

03035-0-032

100

10k

10 100 1k 10k 100k 1M

SAMPLING RATE (SPS)

POWER DISSIPATION (µW)

WARP MODE POWER

PDREF = PDBUF = HIGH

100k

IMPULSE MODE POWER

Figure 32. Power Dissipation vs. Sampling Rate

AD7667

Rev. 0 | Page 21 of 28

CONVERSION CONTROL

Figure 33 shows the detailed timing diagrams of the conversion

process. The AD7667 is controlled by the CNVST signal, which

initiates conversion. Once initiated, it cannot be restarted or

aborted, even by the power-down input PD, until the conversion

is complete. CNVST operates independently of CS and RD.

In Impulse mode, conversions can be automatically initiated. If

CNVST is held LOW when BUSY is LOW, the AD7667 controls

the acquisition phase and automatically initiates a new con-

version. By keeping CNVST LOW, the AD7667 keeps the

conversion process running by itself. It should be noted that the

analog input must be settled when BUSY goes LOW. Also, at

power-up, CNVST should be brought LOW once to initiate the

conversion process. In this mode, the AD7667 can run slightly

faster than the guaranteed 666 kSPS limits in Impulse mode.

This feature does not exist in Warp and Normal modes.

Although CNVST is a digital signal, it should be designed with

special care with fast, clean edges, and levels with minimum

overshoot and undershoot or ringing.

The CNVST trace should be shielded with ground and a low

value serial resistor (e.g., 50 Ω) termination should be added

close to the output of the component that drives this line.

For applications where SNR is critical, the CNVST signal should

have very low jitter. This can be achieved by using a dedicated

oscillator for CNVST generation, or by clocking CNVST with a

high frequency, low jitter clock, as shown in Figure 26.

BUSY

MODE

ACQUIRE CONVERT ACQUIRE CONVERT

03033-0-026

CNVST

Figure 33. Basic Conversion Timing

RESET

DATA

BUSY

03033-0-027

CNVST

Figure 34. RESET Timing

BUSY

DATA

BUS

CS = RD = 0

PREVIOUS CONVERSION DATA NEW DATA

03033-0-028

CNVST

Figure 35. Master Parallel Data Timing for Reading (Continuous Read)

AD7667

Rev. 0 | Page 22 of 28

DIGITAL INTERFACE

The AD7667 has a versatile digital interface; it can be interfaced

with the host system by using either a serial or a parallel

interface. The serial interface is multiplexed on the parallel data

bus. The AD7667 digital interface also accommodates both 3 V

and 5 V logic by simply connecting the OVDD supply pin of the

AD7667 to the host system interface digital supply. Finally, by

using the OB/2C input pin, both twos complement or straight

binary coding can be used.

The two signals, CS and RD, control the interface. CS and RD

have a similar effect because they are OR’d together internally.

When at least one of these signals is HIGH, the interface

outputs are in high impedance. Usually CS allows the selection

of each AD7667 in multicircuit applications and is held low in a

single AD7667 design. RD is generally used to enable the

conversion result on the data bus.

PARALLEL INTERFACE

The AD7667 is configured to use the parallel interface when

SER/PAR is held LOW. The data can be read either after each

conversion, which is during the next acquisition phase, or

during the following conversion, as shown in Figure 36 and

Figure 37, respectively. When the data is read during the

conversion, however, it is recommended that it is read only

during the first half of the conversion phase. This avoids any

potential feedthrough between voltage transients on the digital

interface and the most critical analog conversion circuitry.

The BYTESWAP pin allows a glueless interface to an 8-bit bus.

As shown in Figure 38, the LSB byte is output on D[7:0] and the

MSB is output on D[15:8] when BYTESWAP is LOW. When

BYTESWAP is HIGH, the LSB and MSB bytes are swapped and

the LSB is output on D[15:8] and the MSB is output on D[7:0].

By connecting BYTESWAP to an address line, the 16-bit data

can be read in two bytes on either D[15:8] or D[7:0].

SERIAL INTERFACE

The AD7667 is configured to use the serial interface when

SER/PAR is held HIGH. The AD7667 outputs 16 bits of data,

MSB first, on the SDOUT pin. This data is synchronized with

the 16 clock pulses provided on the SCLK pin. The output data

is valid on both the rising and falling edges of the data clock.

CURRENT

CONVERSION

BUSY

DATA

BUS

03033-0-029

Figure 36. Slave Parallel Data Timing for Reading (Read after Convert)

CONVERSION

BUSY

DATA

BUS

03033-0-030

CNVST,

CS = 0

Figure 37. Slave Parallel Data Timing for Reading (Read during Convert)

BYTESWAP

PINS D[15:8]

PINS D[7:0] HI-Z

HI-Z HIGH BYTE LOW BYTE

LOW BYTE HIGH BYTE HI-Z

HI-Z

03033-0-031

Figure 38. 8-Bit Parallel Interface

AD7667

Rev. 0 | Page 23 of 28

MASTER SERIAL INTERFACE Usually, because the AD7667 is used with a fast throughput, the

Master Read During Conversion mode is the most recommen-

ded serial mode. In this mode, the serial clock and data toggle at

appropriate instants, minimizing potential feedthrough between

digital activity and critical conversion decisions.

Internal Clock

The AD7667 is configured to generate and provide the serial

data clock SCLK when the EXT/INT pin is held LOW. The

AD7667 also generates a SYNC signal to indicate to the host

when the serial data is valid. The serial clock SCLK and the

SYNC signal can be inverted if desired. Depending on the

RDC/SDIN input, the data can be read after each conversion or

during the following conversion. Figure 39 and Figure 40 show

detailed timing diagrams of these two modes.

In Read After Conversion mode, it should be noted that unlike

in other modes, the BUSY signal returns LOW after the 16 data

bits are pulsed out and not at the end of the conversion phase,

which results in a longer BUSY width.

BUSY

SYNC

SCLK

SDOUT

123 141516

D15 D14 D2 D1 D0

RDC/SDIN = 0 INVSCLK = INVSYNC = 0

03033-0-032

CNVST

CS, RD

EXT/INT = 0

Figure 39. Master Serial Data Timing for Reading (Read after Convert)

EXT/INT = 0 RDC/SDIN = 1 INVSCLK = INVSYNC = 0

D15 D14 D2 D1 D0X

12 3 141516

BUSY

SYNC

SCLK

SDOUT

03033-0-033

CNVST

CS, RD

Figure 40. Master Serial Data Timing for Reading (Read Previous Conversion during Convert)

AD7667

Rev. 0 | Page 24 of 28

SLAVE SERIAL INTERFACE

External Clock

The AD7667 is configured to accept an externally supplied

serial data clock on the SCLK pin when the EXT/INT pin is

held HIGH. In this mode, several methods can be used to read

the data. The external serial clock is gated by CS. When CS and

RD are both LOW, the data can be read after each conversion or

during the following conversion. The external clock can be

either a continuous or a discontinuous clock. A discontinuous

clock can be either normally HIGH or normally LOW when

inactive. Figure 41 and Figure 42 show the detailed timing

diagrams of these methods. Usually, because the AD7667 has a

longer acquisition phase than conversion phase, the data are

read immediately after conversion.

While the AD7667 is performing a bit decision, it is important

that voltage transients be avoided on digital input/output pins

or degradation of the conversion result could occur. This is

particularly important during the second half of the conversion

phase because the AD7667 provides error correction circuitry

that can correct for an improper bit decision made during the

first half of the conversion phase. For this reason, it is

recommended that when an external clock is being provided, it

is a discontinuous clock that is toggling only when BUSY is

LOW, or, more importantly, that it does not transition during

the latter half of BUSY HIGH.

SCLK

SDOUT D15 D14 D1 D0

D13

X15 X14 X13 X1 X0 Y15 Y14

BUSY

SDIN

INVSCLK = 0

X15 X14

1 2 3 14151617 18

03033-0-034

EXT/INT = 1

RD RD

= 0

Figure 41. Slave Serial Data Timing for Reading (Read after Convert)

DOUT

SCLK

D1 D0

XD15 D14 D13

123 141516

t3t35

t36 t37

t31 t32

t16

BUSY

EXT/INT = 1 INVSCLK = 0

03033-0-035

CNVST

CS RD = 0

Figure 42. Slave Serial Data Timing for Reading (Read Previous Conversion during Convert)

AD7667

Rev. 0 | Page 25 of 28

External Discontinuous Clock Data Read After

Conversion

Though the maximum throughput cannot be achieved using

this mode, it is the most recommended of the serial slave

modes. Figure 41 shows the detailed timing diagrams of this

method. After a conversion is complete, indicated by BUSY

returning LOW, the conversion result can be read while both CS

and RD are LOW. Data is shifted out MSB first with 16 clock

pulses and is valid on the rising and falling edges of the clock.

Among the advantages of this method is the fact that conver-

sion performance is not degraded because there are no voltage

transients on the digital interface during the conversion process.

Another advantage is the ability to read the data at any speed up

to 40 MHz, which accommodates both the slow digital host

interface and the fastest serial reading.

Finally, in this mode only, the AD7667 provides a daisy-chain

feature using the RDC/SDIN pin for cascading multiple con-

verters together. This feature is useful for reducing component

count and wiring connections when desired, as, for instance, in

isolated multiconverter applications.

An example of the concatenation of two devices is shown in

Figure 43. Simultaneous sampling is possible by using a

common CNVST signal. It should be noted that the RDC/SDIN

input is latched on the opposite edge of SCLK of the one used to

shift out the data on SDOUT. Therefore, the MSB of the

“upstream” converter just follows the LSB of the “downstream”

converter on the next SCLK cycle.

SCLK

SDOUTRDC/SDIN

BUSYBUSY

DATA

OUT

AD7667

(DOWNSTREAM)

BUSY

OUT

SCLK

AD7667

(UPSTREAM)

RDC/SDIN SDOUT

SCLK IN

CNVST IN

03035-0-036

CNVST

CS IN

Figure 43. Two AD7667s in a Daisy-Chain Configuration

External Clock Data Read During Conversion

Figure 42 shows the detailed timing diagrams of this method.

During a conversion, while both CS and RD are LOW, the result

of the previous conversion can be read. The data is shifted out

MSB first with 16 clock pulses, and is valid on both the rising

and falling edges of the clock. The 16 bits must be read before

the current conversion is complete; otherwise, RDERROR is

pulsed HIGH and can be used to interrupt the host interface to

prevent incomplete data reading. There is no daisy-chain feature

in this mode and the RDC/SDIN input should always be tied

either HIGH or LOW.

To reduce performance degradation due to digital activity, a fast

discontinuous clock (at least 18 MHz when Impulse mode is

used, 25 MHz when Normal mode is used, or 40 MHz when

Warp mode is used) is recommended to ensure that all the bits

are read during the first half of the conversion phase. It is also

possible to begin to read data after conversion and continue to

read the last bits after a new conversion has been initiated. This

allows the use of a slower clock speed like 14 MHz in Impulse

mode, 18 MHz in Normal mode, and 25 MHz in Warp mode.

AD7667

Rev. 0 | Page 26 of 28

MICROPROCESSOR INTERFACING

The AD7667 is ideally suited for traditional dc measurement

applications supporting a microprocessor, and for ac signal

processing applications interfacing to a digital signal processor.

The AD7667 is designed to interface either with a parallel 8-bit

or 16-bit wide interface, or with a general-purpose serial port or

I/O ports on a microcontroller. A variety of external buffers can

be used with the AD7667 to prevent digital noise from coupling

into the ADC. The following section discusses the use of an

AD7667 with an ADSP-219x SPI equipped DSP.

SPI Interface (ADSP-219x)

Figure 44 shows an interface diagram between the AD7667 and

the SPI equipped ADSP-219x. To accommodate the slower

speed of the DSP, the AD7667 acts as a slave device and data

must be read after conversion. This mode also allows the daisy-

chain feature. The convert command can be initiated in

response to an internal timer interrupt. The reading process can

be initiated in response to the end-of-conversion signal (BUSY

going LOW) using an interrupt line of the DSP. The serial inter-

face (SPI) on the ADSP-219x is configured for master mode—

(MSTR) = 1, Clock Polarity bit (CPOL) = 0, Clock Phase bit

(CPHA) = 1, and SPI Interrupt Enable (TIMOD) = 00—by

writing to the SPI control register (SPICLTx). To meet all timing

requirements, the SPI clock should be limited to 17 Mbps,

which allows it to read an ADC result in less than 1 µs. When a

higher sampling rate is desired, use of one of the parallel

interface modes is recommended.

AD7667*

ADSP-219x*

SER/PAR

PFx

MISOx

SCKx

PFx or TFSx

BUSY

SDOUT

SCLK

CNVST

EXT/INT

INVSCLK

DVDD

*ADDITIONAL PINS OMITTED FOR CLARITY

SPIxSEL (PFx)

03035-0-037

Figure 44. Interfacing the AD7667 to an SPI Interface

AD7667

Rev. 0 | Page 27 of 28

APPLICATION HINTS

BIPOLAR AND WIDER INPUT RANGES

In some applications, it is desirable to use a bipolar or wider

analog input range such as ±10 V, ±5 V, or 0 V to 5 V. Although

the AD7667 has only one unipolar range, simple modifications

of input driver circuitry allow bipolar and wider input ranges to

be used without any performance degradation. Figure 45 shows

a connection diagram that allows this. Component values

required and resulting full-scale ranges are shown in Table 8.

When desired, accurate gain and offset can be calibrated by

acquiring a ground and voltage reference using an analog

multiplexer (U2), as shown in Figure 45.

NALO

INPUT R2

R3 R4 100nF

REF

INGND

REF

REFGND

AD7667

03035-0-038

15Ω

2.7nF

Figure 45. Using the AD7667 in 16-Bit Bipolar and/or Wider Input Ranges

Table 8. Component Values and Input Ranges

Input Range R1 (Ω) R2 (kΩ) R3 (kΩ) R4 (kΩ)

±10 V 500 4 2.5 2

±5 V 500 2 2.5 1.67

0 V to –5 V 500 1 None 0

LAYOUT

The AD7667 has very good immunity to noise on the power

supplies. However, care should still be taken with regard to

grounding layout.

The printed circuit board that houses the AD7667 should be

designed so the analog and digital sections are separated and

confined to certain areas of the board. This facilitates the use of

ground planes that can be separated easily. Digital and analog

ground planes should be joined in only one place, preferably

underneath the AD7667, or as close as possible to the AD7667.

If the AD7667 is in a system where multiple devices require

analog-to-digital ground connections, the connection should

still be made at one point only, a star ground point that should

be established as close as possible to the AD7667.

Running digital lines under the device should be avoided since

these couple noise onto the die. The analog ground plane

should be allowed to run under the AD7667 to avoid noise

coupling. Fast switching signals like CNVST or clocks should be

shielded with digital ground to avoid radiating noise to other

sections of the board, and should never run near analog signal

paths. Crossover of digital and analog signals should be avoided.

Traces on different but close layers of the board should run at

right angles to each other. This will reduce the effect of crosstalk

through the board.

The power supply lines to the AD7667 should use as large a

trace as possible to provide low impedance paths and reduce the

effect of glitches on the power supply lines. Good decoupling is

also important to lower the supply’s impedance presented to the

AD7667 and to reduce the magnitude of the supply spikes.

Decoupling ceramic capacitors, typically 100 nF, should be

placed on each power supply pin—AVDD, DVDD, and

OVDD—close to, and ideally right up against these pins and

their corresponding ground pins. Additionally, low ESR 10 µF

capacitors should be located near the ADC to further reduce

low frequency ripple.

The DVDD supply of the AD7667 can be a separate supply or

can come from the analog supply AVDD or the digital interface

supply OVDD. When the system digital supply is noisy or when

fast switching digital signals are present, if no separate supply is

available, the user should connect DVDD to AVDD through an

RC filter (see Figure 26) and the system supply to OVDD and

the remaining digital circuitry. When DVDD is powered from

the system supply, it is useful to insert a bead to further reduce

high frequency spikes.

The AD7667 has five different ground pins: INGND, REFGND,

AGND, DGND, and OGND. INGND is used to sense the analog

input signal. REFGND senses the reference voltage and, because

it carries pulsed currents, should be a low impedance return to

the reference. AGND is the ground to which most internal ADC

analog signals are referenced; it must be connected with the

least resistance to the analog ground plane. DGND must be tied

to the analog or digital ground plane depending on the config-

uration. OGND is connected to the digital system ground.

EVALUATING THE AD7667’S PERFORMANCE

A recommended layout for the AD7667 is outlined in the

EVAL-AD7667 evaluation board for the AD7667. The

evaluation board package includes a fully assembled and tested

evaluation board, documentation, and software for controlling

the board from a PC via the EVAL-CONTROL BRD2.

AD7667

Rev. 0 | Page 28 of 28

OUTLINE DIMENSIONS

TOP VIEW

(PINS DOWN)

12 13 25

0.27

0.22

0.17

0.50

BSC

7.00

BSC SQ

SEATING

PLANE

1.60

MAX

0.75

0.60

0.45

VIEW A

9.00 BSC

PIN 1

0.20

0.09

1.45

1.40

1.35

0.08 MAX

COPLANARITY

VIEW A

ROTATED 90° CCW

SEATING

PLANE

10°

6°

2°

7°

3.5°

0°

0.15

0.05

COMPLIANT TO JEDEC STANDARDS MS-026BBC

Figure 46. 48-Lead Quad Flatpack (LQFP) [ST-48]

Dimensions shown in millimeters

PIN 1

INDICATOR

TOP

BSC SQ

7.00

BSC SQ

BOTTOM

5.10 SQ

4.95

0.50

0.40

0.30

0.23

0.18

0.50 BSC



12° MAX 0.80 MAX

0.65 TYP

1.00

0.85

0.80

5.50

REF

0.05 MAX

0.02 NOM

0.60 MAX

0.60 MAX PIN 1

INDICATOR

COPLANARITY

0.08

SEATING

PLANE

PADDLE CONNECTED TO AGND.

THIS CONNECTION IS NOT

REQUIRED TO MEET THE

ELECTRICAL PERFORMANCES

0.25 MIN

0.20 REF

COMPLIANT TO JEDEC STANDARDS MO-220-VKKD-2

Figure 47. 48-Lead Frame Chip Scale Package (LFCSP) [CP-48]

Dimensions shown in millimeters

ORDERING GUIDE

Model Temperature Range Package Description Package Option

AD7667AST –40°C to +85°C Quad Flatpack (LQFP) ST-48

AD7667ASTRL –40°C to +85°C Quad Flatpack (LQFP) ST-48

AD7667ACP –40°C to +85°C Lead Frame Chip Scale (LFCSP) CP-48

AD7667ACPRL –40°C to +85°C Lead Frame Chip Scale (LFCSP) CP-48

EVAL-AD7667CB1 Evaluation Board

EVAL-CONTROL BRD22 Controller Board

1This board can be used as a standalone evaluation board or in conjunction with the EVAL-CONTROL BRD2 for evaluation/demonstration purposes.

2This board allows a PC to control and communicate with all Analog Devices evaluation boards ending in the CB designators.

registered trademarks are the property of their respective owners.

D03035–0–1/04(0)