DPA425RN-TL - Power Integrations, Inc.

Figure 1. Typical Forward Converter Application.

DPA423-426

DPA-Switch® Family

Highly Integrated DC-DC Converter ICs

for Distributed Power Architectures

December 2004

Product Highlights

Highly Integrated Solution

• Eliminates 20-50 external components–saves space, cost

• Integrates 220 V high frequency MOSFET, PWM control

• Lower cost plastic DIP surface mount (G package) and

through-hole (P package) options for designs ≤35 W

Superior Performance and Flexibility

• Eliminates all external current sensing circuitry

• Built-in auto-restart for output overload/open loop protection

• Pin selectable 300/400 kHz ﬁxed frequency

• Wide input (line) voltage range: 16-75 VDC

• Source connected tab reduces EMI

• Line under-voltage (UV) detection: meets ETSI standards

• Line overvoltage (OV) shutdown protection

• Low cost synchronous rectiﬁcation: line UV/OV shut-down

limits gate drive voltage range from transformer winding

• Fully integrated soft-start for minimum stress/overshoot

• Externally programmable current limit

• Supports forward or ﬂyback topology

• Cycle skipping: regulation to zero load without pre-load

• Hysteretic thermal shutdown for automatic fault recovery

EcoSmart®- Energy Efﬁcient

• Extremely low consumption at no load

• Cycle skipping at light load for high standby efﬁciency

Applications

• Telco central ofﬁce equipment: xDSL, ISDN, PABX, etc.

• Distributed power architectures (24 V/48 V bus, etc.)

• Digital feature phones, VoIP phones, PoE

• Industrial controls (24 V/48 V)

Description

The DPA-Switch IC family introduces a highly integrated

solution for DC-DC conversion applications in the 16-75 VDC

input range.

DPA-Switch uses the same proven topology as TOPSwitch,

cost effectively integrating the high voltage power

MOSFET, PWM control, fault protection and other control

circuitry onto a single CMOS chip. High performance

features are enabled with three user conﬁgurable pins. The

following transparent or built-in features are also provided:

soft-start, cycle skipping down to zero load and hysteretic

thermal shutdown. In addition, all critical parameters

(i.e. current limit, frequency, PWM gain) have tight temperature and

absolute tolerance, to simplify design and optimize system cost.

Table 1. Notes: 1. Maximum output power is limited by device internal

current limit. 2. See Applications Considerations section for complete

description of assumptions and for output powers with other input

voltage ranges. 3. For device dissipation of 1.5 W or below, use P or G

packages. Device dissipation above 1.5 W is possible only with R package.

4. Packages: P: DIP-8, G: SMD-8, R: TO-263-7C. For lead-free package

options, see Part Ordering Information. 5. Available in R package only.

6. Due to higher switching losses, the DPA425 may not deliver additional

power compared to a smaller device.

PI-2770-032002

DPA-Switch

VIN

CONTROL

SENSE

CIRCUIT

RESET/

CLAMP

CIRCUIT

OUTPUT POWER TABLE

36-75 INPUT RANGE (FORWARD)2

Total Device

Disipation3

PRODUCT4

0.5 W 1 W 2.5 W 4 W 6 W

Max

Power

Output1

DPA423 12 W 16 W - - - 18 W

DPA424 16 W 23 W 35 W - - 35 W

DPA425 23 W 32 W 50 W 62 W 70 W

DPA426525 W 35 W 55 W 70 W 83 W 100 W

36-75 INPUT RANGE (FLYBACK)2

Total Device

Disipation3

PRODUCT4

0.5 W 0.75 W 1 W 1.5 W

Max

Power

Output1

DPA423 9 W 13 W - - 13 W

DPA424 10 W 14.5 W 18 W 24 W 26 W

DPA425 -6-6-625.5 W 52 W

DPA423-426

12/04

Section List

Functional Block Diagram ....................................................................................................................................... 3

Pin Functional Description ...................................................................................................................................... 3

DPA-Switch Family Functional Description ............................................................................................................ 4

CONTROL (C) Pin Operation .................................................................................................................................. 4

Oscillator and Switching Frequency ....................................................................................................................... 5

Pulse Width Modulator & Maximum Duty Cycle ..................................................................................................... 6

Minimum Duty Cycle and Cycle Skipping .............................................................................................................. 6

Error Ampliﬁer ......................................................................................................................................................... 6

On-chip Current Limit with External Programmability ............................................................................................. 6

Line Under-Voltage Detection (UV) ......................................................................................................................... 6

Line Overvoltage Shutdown (OV) ........................................................................................................................... 7

Line Feed-Forward with DCMAX Reduction .............................................................................................................. 7

Remote ON/OFF ..................................................................................................................................................... 7

Synchronization ...................................................................................................................................................... 8

Soft-Start ................................................................................................................................................................. 8

Shutdown/Auto-Restart ........................................................................................................................................... 8

Hysteretic Over-Temperature Protection ................................................................................................................. 8

Bandgap Reference ............................................................................................................................................... 8

High-Voltage Bias Current Source .......................................................................................................................... 8

Using Feature Pins ..................................................................................................................................................... 9

FREQUENCY (F) Pin Operation .............................................................................................................................. 9

LINE-SENSE (L) Pin Operation ............................................................................................................................... 9

EXTERNAL CURRENT LIMIT (X) Pin Operation ...................................................................................................... 9

Typical Uses of FREQUENCY (F) Pin ...................................................................................................................... 12

Typical Uses of LINE-SENSE (L) and EXTERNAL CURRENT LIMIT (X) Pins ...................................................... 12

Application Examples .............................................................................................................................................. 15

Key Application Considerations ............................................................................................................................. 17

DPA-Switch Design Considerations ...................................................................................................................... 17

DPA-Switch Layout Considerations ...................................................................................................................... 18

Quick Design Checklist ........................................................................................................................................ 19

Design Tools ......................................................................................................................................................... 19

Product Speciﬁcations and Test Conditions ......................................................................................................... 21

Typical Performance Characteristics .................................................................................................................. 27

Part Ordering Information ....................................................................................................................................... 31

Package Outlines .................................................................................................................................................... 32

DPA423-426

12/04 3

Figure 2. Functional Block Diagram.

PI-2760-070501

SHUTDOWN/

AUTO-RESTART

PWM

COMPARATOR

CLOCK

SAW

300/400 kHz

CONTROLLED

TURN-ON

GATE DRIVER

CURRENT LIMIT

COMPARATOR

INTERNAL UV

COMPARATOR

INTERNAL

SUPPLY

5.8 V

4.8 V

SOURCE (S)

DMAX

STOP SOFT-

START

CONTROL (C)

LINE-SENSE (L)

EXTERNAL

CURRENT LIMIT (X)

FREQUENCY (F)

5.8 V

1 V

IFB

LEADING

EDGE

BLANKING

÷ 8

HYSTERETIC

THERMAL

SHUTDOWN

SHUNT REGULATOR/

ERROR AMPLIFIER

DRAIN (D)

ON/OFF

SOFT

START

DCMAX

VBG

DCMAX

VBG + VT

OV/UV

VI (LIMIT)

CURRENT

LIMIT

ADJUST

LINE

SENSE

SOFT START

CYCLE

SKIPPING

STOP LOGIC

OSCILLATOR

Figure 3. Pin Conﬁguration (top view).

PI-3822-041304

Tab Internally Connected to

SOURCE Pin (See layout considerations)

R Package

(TO-263-7C)

1 2 3 4 5 7

C L X S F D

P Package (DIP-8)

G Package (SMD-8)

Pin Functional Description

DRAIN (D) Pin:

High voltage power MOSFET drain output. The internal startup

bias current is drawn from this pin through a switched high-

voltage current source. Internal current limit sense point for

drain current.

CONTROL (C) Pin:

Error ampliﬁer and feedback current input pin for duty cycle

control. Internal shunt regulator connection to provide

internal bias current during normal operation. It is also used

as the connection point for the supply bypass and auto-restart/

compensation capacitor.

LINE-SENSE (L) Pin:

Input pin for overvoltage (OV), under-voltage (UV) lock out,

line feed-forward with the maximum duty cycle (DCMAX)

reduction, remote ON/OFF and synchronization. A connection

to SOURCE pin disables all functions on this pin.

EXTERNAL CURRENT LIMIT (X) Pin:

Input pin for external current limit adjustment and remote

ON/OFF. A connection to SOURCE pin disables all functions

on this pin.

FREQUENCY (F) Pin:

Input pin for selecting switching frequency: 400 kHz if connected

to SOURCE pin and 300 kHz if connected to CONTROL

pin.

SOURCE (S) Pin:

Output MOSFET source connection for the power return.

Primary side control circuit common and reference point.

DPA423-426

12/04

DPA-Switch Family Functional

Description

DPA-Switch is an integrated switched mode power supply

chip that converts a current at the control input to a duty cycle

at the open drain output of a high voltage power MOSFET.

During normal operation the duty cycle of the power MOSFET

decreases linearly with increasing CONTROL pin current as

shown in Figure 4. A patented high-voltage CMOS technology

allows both the high-voltage power MOSFET and all the low

voltage control circuitry to be cost effectively integrated onto

a single monolithic chip.

In addition to the standard TOPSwitch features, such as the

high-voltage start-up, the cycle-by-cycle current limiting, loop

compensation circuitry, auto-restart and thermal shutdown,

DPA-Switch also offers many advanced features that reduce

system cost and increase power supply performance and design

ﬂexibility. Following is a summary of the advanced features:

1. A fully integrated 5 ms soft-start limits peak currents and

voltages during start-up and reduces or eliminates output

overshoot in most applications.

2. A 75% maximum duty cycle (DCMAX) together with the

line feed-forward with DCMAX reduction feature makes

DPA-Switch well suited for both ﬂyback and forward

topologies.

3. High switching frequency (400 kHz/300 kHz, pin selectable)

allows the use of smaller size transformers and offers high

bandwidth for power supply control loop.

4. Cycle skipping operation at light load minimizes standby

power consumption (typically <10 mA input current).

5. Line under-voltage ensures glitch free operations at both

power-up and power-down and is tightly toleranced over

process and temperature to meet system level requirements

common in DC to DC converters (e.g. ETSI).

6. Line overvoltage protects DPA-Switch against excessive

input voltage and line surge.

7. External current limit adjustment allows the setting of the

current limit externally to a lower level near the operating

peak current and, if desired, further adjusts the level

gradually as line voltage rises. This makes possible an ideal

implementation of overload power limiting.

8. Synchronization function allows the synchronization of

DPA-Switch operation to an external lower frequency.

9. Remote ON/OFF feature permits DPA-Switch based power

supplies to be easily switched on/off using logic signals.

Maximum input current consumption is 2 mA in remote

OFF.

10. Hysteretic over-temperature shutdown provides automatic

recovery from thermal fault.

11. Tight absolute tolerances and small temperature variations

on switching frequency, current limit, and under-voltage

lock out threshold (UV).

Three pins, LINE-SENSE (L), EXTERNAL CURRENT LIMIT

(X) and FREQUENCY (F), are used to implement all the pin

controllable features. A resistor from the LINE-SENSE pin to DC

input bus implements line UV, line OV and line feed-forward with

DCMAX reduction. A resistor from the EXTERNAL CURRENT

LIMIT pin to the SOURCE pin sets current limit externally. In

addition, remote ON/OFF may be implemented through either

the LINE-SENSE pin or the EXTERNAL CURRENT LIMIT

pin depending on the polarity of the logic signal available as

well as other system speciﬁc considerations. Shorting both the

LINE-SENSE and the EXTERNAL CURRENT LIMIT pins to

the SOURCE pin disables line OV, line UV, line feed-forward

with DCMAX reduction, external current limit, remote ON/OFF

and synchronization. The FREQUENCY pin sets the switching

frequency to 400 kHz if connected to the SOURCE pin, or

300 kHz if connected to the CONTROL pin. This pin should

not be left open. Please refer to “Using Feature Pins” section for

detailed information regarding the proper use of those pins.

CONTROL Pin Operation

The CONTROL pin is a low impedance node that is capable

of receiving a combined supply and feedback current. During

normal operation, a shunt regulator is used to separate the

feedback signal from the supply current. CONTROL pin voltage

VC is the supply voltage for the control circuitry including the

MOSFET gate driver. An external bypass capacitor closely

connected between the CONTROL and SOURCE pins is required

to supply the instantaneous gate drive current. The total amount

of capacitance connected to this pin also sets the auto-restart

timing as well as control loop compensation.

When the DC input voltage is applied to the DRAIN pin during

start-up, the MOSFET is initially off, and the CONTROL

pin capacitor is charged through the switched high voltage

current source connected internally between the DRAIN and

CONTROL pins. When the CONTROL pin voltage VC reaches

Figure 4. Relationship of Duty Cycle to CONTROL Pin Current.

PI-2761-112102

Duty Cycle (%)

IC (mA)

IL = 115 µA

IL < IL(DC) IC (SKIP)

Slope = PWM Gain

ICD1

Auto-restart

DPA423-426

12/04 5

approximately 5.8 V, the control circuitry is activated and the

soft-start begins. The soft-start circuit gradually increases

the duty cycle of the MOSFET from zero to the maximum

value over approximately 5 ms. The high voltage current

source is turned off at the end of the soft-start. If no external

feedback/supply current is fed into the CONTROL pin by the

end of the soft-start, the CONTROL pin will start discharging

in response to the supply current drawn by the control circuitry

and the gate current of the switching MOSFET driver. If the

power supply is designed properly, and no fault condition such

as open loop or overloaded output exists, the feedback loop

will close, providing external CONTROL pin current, before

the CONTROL pin voltage has had a chance to discharge to

the lower threshold voltage of approximately 4.8 V (internal

supply under-voltage lockout threshold). When the externally

fed current charges the CONTROL pin to the shunt regulator

voltage of 5.8 V, current in excess of the consumption of the

chip is shunted to SOURCE through resistor RE as shown in

Figure 2. This current ﬂowing through RE controls the duty cycle

of the power MOSFET to provide closed loop regulation. The

shunt regulator has a ﬁnite low output impedance ZC that sets

the gain of the error ampliﬁer when used in a primary feedback

conﬁguration. The dynamic impedance ZC of the CONTROL

pin together with the external CONTROL pin capacitance sets

the dominant pole for the control loop.

When a fault condition such as an open loop or overloaded output

prevents the ﬂow of an external current into the CONTROL

pin, the capacitor on the CONTROL pin discharges towards

4.8 V. At 4.8 V auto-restart is activated which turns the output

MOSFET off and puts the control circuitry in a low current

standby mode. The high-voltage current source turns on and

charges the external capacitance again. A hysteretic internal

supply under-voltage comparator keeps VC within a window

of typically 4.8 V to 5.8 V by turning the high-voltage current

source on and off as shown in Figure 5. The auto-restart circuit

has a divide-by-8 counter that prevents the output MOSFET

from turning on again until eight discharge/charge cycles have

elapsed. This is accomplished by enabling the output MOSFET

only when the divide-by-8 counter reaches full count (S7).

The counter effectively limits DPA-Switch power dissipation

as well as the maximum power delivered to the power supply

output by reducing the auto-restart duty cycle to typically 4%.

Auto-restart mode continues until output voltage regulation is

again achieved through closure of the feedback loop.

Oscillator and Switching Frequency

The internal oscillator linearly charges and discharges an internal

capacitance between two voltage levels to create a sawtooth

waveform for the pulse width modulator. The oscillator sets

both the pulse width modulator latch and the current limit latch

at the beginning of each cycle.

The nominal switching frequency of 400 kHz was chosen to

minimize the transformer size and to allow faster power supply

loop response. The FREQUENCY pin, when shorted to the

CONTROL pin, lowers the switching frequency to 300 kHz,

which may be preferable in some applications such as those

employing secondary synchronous rectiﬁcation. Otherwise, the

FREQUENCY pin should be connected to the SOURCE pin

for the default 400 kHz.

Figure 5. Typical Waveforms for (1) Power Up (2) Normal Operation (3) Auto-restart (4) Power Down.

PI-3867-050602

S1 S2 S6 S7 S1 S2 S6 S7S0 S1 S7

S0 S0 5.8 V

4.8 V

0 V

VLINE

VDRAIN

VOUT

Note: S0 through S7 are the output states of the auto-restart counter

1234

0 V

S6 S7

VUV

DPA423-426

12/04

Pulse Width Modulator and Maximum Duty Cycle

The pulse width modulator implements voltage mode control

by driving the output MOSFET with a duty cycle inversely

proportional to the current into the CONTROL pin that

is in excess of the internal supply current of the chip (see

Figure 4). The excess current is the feedback error signal that

appears across RE (see Figure 2). This signal is ﬁltered by an RC

network with a typical corner frequency of 30 kHz to reduce the

effect of switching noise in the chip supply current generated by

the MOSFET gate driver. The ﬁltered error signal is compared

with the internal oscillator sawtooth waveform to generate the

duty cycle waveform. As the control current increases, the duty

cycle decreases. A clock signal from the oscillator sets a latch

that turns on the output MOSFET. The pulse width modulator

resets the latch, turning off the output MOSFET. Note that a

minimum current must be driven into the CONTROL pin before

the duty cycle begins to change.

The maximum duty cycle, DCMAX is set at a default maximum

value of 75% (typical). However, by connecting the

LINE-SENSE to the DC input bus through a resistor

with appropriate value, the maximum duty cycle can be

made to decrease from 75% to 33% (typical) as shown in

Figure 7 when input line voltage increases (see Line Feed-

Forward with DCMAX Reduction).

Minimum Duty Cycle and Cycle Skipping

To maintain power supply output regulation, the pulse width

modulator reduces duty cycle as the load at the power supply

output decreases. This reduction in duty cycle is proportional to

the current ﬂowing into the CONTROL pin. As the CONTROL

pin current increases, the duty cycle reduces linearly towards a

minimum value speciﬁed as minimum duty cycle, DCMIN. After

reaching DCMIN, if CONTROL pin current is increased further

by approximately 2 mA, the pulse width modulator will force

the duty cycle from DCMIN to zero in a discrete step (refer to

Figure 4). This feature allows a power supply to operate in a

cycle skipping mode when the load consumes less power than

the DPA-Switch delivers at minimum duty cycle, DCMIN. No

additional control is needed for the transition between normal

operation and cycle skipping. As the load increases or decreases,

the power supply automatically switches between normal and

cycle skipping mode as necessary.

Cycle skipping may be avoided, if so desired, by connecting

a minimum load at the power supply output such that the duty

cycle remains at a level higher than DCMIN at all times.

Error Ampliﬁer

The shunt regulator can also perform the function of an error

ampliﬁer in primary side feedback applications. The shunt

regulator voltage is accurately derived from a temperature-

compensated bandgap reference. The gain of the error ampliﬁer is

set by the CONTROL pin dynamic impedance. The CONTROL

pin clamps external circuit signals to the VC voltage level.

The CONTROL pin current in excess of the supply current

is separated by the shunt regulator and ﬂows through RE as a

voltage error signal.

On-chip Current Limit with External Programmability

The cycle-by-cycle peak drain current limit circuit uses the

output MOSFET ON-resistance as a sense resistor. A current

limit comparator compares the output MOSFET on-state

drain to source voltage, VDS(ON) with a threshold voltage. At

the current limit, VDS(ON) exceeds the threshold voltage and the

MOSFET is turned off until the start of the next clock cycle.

The current limit comparator threshold voltage is temperature

compensated to minimize the variation of the current limit due

to temperature related changes in RDS(ON) of the output MOSFET.

The default current limit of DPA-Switch is preset internally.

However, with a resistor connected between EXTERNAL

CURRENT LIMIT pin and SOURCE pin, the current limit can

be programmed externally to a lower level between 25% and

100% of the default current limit. Please refer to the graphs

in the Typical Performance Characteristics section for the

selection of the resistor value. By setting current limit low,

a larger DPA-Switch than necessary for the power required

can be used to take advantage of the lower RDS(ON) for higher

efﬁciency/smaller heat sinking requirements. With a second

resistor connected between the EXTERNAL CURRENT

LIMIT pin and the DC input bus, the current limit is reduced

with increasing line voltage, allowing a true power limiting

operation against line variation to be implemented in a ﬂyback

conﬁguration.

The leading edge blanking circuit inhibits the current limit

comparator for a short time after the output MOSFET is turned

on. The leading edge blanking time has been set so that, if a

power supply is designed properly, current spikes caused by

primary-side capacitance and secondary-side rectiﬁer reverse

recovery time should not cause premature termination of the

switching pulse.

The current limit after the leading edge blanking time is as shown

in Figure 31. To avoid triggering the current limit in normal

operation, the drain current waveform should stay within the

envelope shown.

Line Under-Voltage Detection (UV)

At power up, UV keeps DPA-Switch off until the input line

voltage reaches the under voltage upper threshold. At power

down, UV holds DPA-Switch on until the input voltage falls

below the under voltage lower threshold. A single resistor

connected from the LINE-SENSE pin to the DC input bus sets

UV upper and lower thresholds. To avoid false triggering by

noise, a hysteresis is implemented which sets the UV lower

threshold typically at 94% of the UV upper threshold. If the UV

lower threshold is reached during operation without the power

supply losing regulation and the condition stays longer than

10 µs (typical), the device will turn off and stay off until the

DPA423-426

12/04 7

Figure 6. Synchronization Timing Diagram.

PI-2762-070501

Oscillator

(SAW)

DMAX

2 V

0 V

tON

fSYNC ≥ 128 kHz; tOFF ≤ 7.7 µs; 120 ns ≤ tON ≤ 2250 ns for fOSC = 400 kHz

≤ 3080 ns for fOSC = 300 kHz

tOFF

ON OFF SYNC

Time

UV upper threshold has been reached again. Then, a soft-start

will be initiated the next time CONTROL pin voltage reaches

5.8.V. If the power supply loses regulation before reaching the

UV lower threshold, the device will enter auto-restart. At the end

of each auto-restart cycle (S7), the UV comparator is enabled. If

the UV upper threshold is not exceeded, the MOSFET will be

disabled during the next cycle (see Figure 5). The UV feature

can be disabled independent of OV feature.

Line Overvoltage Shutdown (OV)

The same resistor used for UV also sets an overvoltage

threshold which, once exceeded, will force the DPA-Switch

output into the off-state within one switching cycle. The ratio

of OV and UV thresholds is preset at 2.7 as can be seen in

Figure 7. When the MOSFET is off, the input voltage surge

capability is increased to the voltage rating of the MOSFET

(220 V), due to the absence of the reﬂected voltage and leakage

spikes on the drain. A small amount of hysteresis is provided on

the OV threshold to prevent noise triggering. The OV feature

can be disabled independent of the UV feature as shown in

Figure 13.

Line Feed-Forward with DCMAX Reduction

The same resistor used for UV and OV also implements line

voltage feed-forward that minimizes output line ripple and

reduces power supply output sensitivity to line transients.

This feed-forward operation is illustrated in Figure 4 by the

different values of IL. Note that for the same CONTROL pin

current, higher line voltage results in smaller operating duty

cycle. As an added feature, the maximum duty cycle DCMAX

is also reduced from 75% (typical) at a voltage slightly higher

than the UV threshold to 33% (typical) at the OV threshold

(see Figures 4, 7). Limiting DCMAX at higher line voltages

helps prevent transformer saturation due to large load transients

in forward converter applications. DCMAX of 33% at the OV

threshold was chosen to ensure that the power capability of

the DPA-Switch is not restricted by this feature under normal

operation.

Remote ON/OFF

Remote ON/OFF control describes operation where the IC is

turned on or off for long periods as opposed to the cycle-by-

cycle on/off control, which is described in the Synchronization

section below.

DPA-Switch can be turned on or off by controlling the current into

the LINE-SENSE pin or out from the EXTERNAL CURRENT

LIMIT pin (see Figure 7). This allows easy implementation of

remote ON/OFF control of DPA-Switch in several different

ways. A transistor or an optocoupler output connected between

the EXTERNAL CURRENT LIMIT pin and the SOURCE pin

implements this function with “active-on” (Figures 17, 19 and 21)

while a transistor or an optocoupler output connected between

the LINE-SENSE pin and the CONTROL pin implements the

function with “active-off” (Figures 18, 20 and 22).

When a signal is received at the LINE-SENSE pin or the

EXTERNAL CURRENT LIMIT pin to disable the output

through any of the pin functions such as OV, UV and remote

ON/OFF, DPA-Switch always completes its current switching

cycle before the output is forced off. The internal oscillator is

stopped at the end of the current cycle and stays there as long

as the disable signal exists. When the signal at the above pins

changes state from disable to enable, the internal oscillator

starts the next switching cycle.

The remote ON/OFF feature can be used as a standby or power

switch to turn off the DPA-Switch and keep it in a very low

power consumption state for indeﬁnitely long periods. If the

DPA-Switch is held in remote-off state for longer than 10 µs

DPA423-426

12/04

(typical), the CONTROL pin goes into the hysteretic mode

of operation. In this mode, the CONTROL pin goes through

alternate charge and discharge cycles between 4.8 V and

5.8 V (see CONTROL Pin Operation section above) and the IC

runs entirely off the high voltage DC input, but with very low

power consumption (30 mW typical at 48 V with LINE-SENSE

and EXTERNAL CURRENT LIMIT pins open). When the

DPA-Switch is remotely turned on after entering this mode,

it will initiate a normal start-up sequence with soft-start the

next time the CONTROL pin reaches 5.8 V. In the worst case,

the delay from remote on to start-up can be equal to the full

discharge/charge cycle time of the CONTROL pin, which is

approximately 36 ms for a 22 µF CONTROL pin capacitor. This

reduced-consumption remote-off mode can eliminate expensive

and unreliable in-line mechanical switches. It also allows for

microprocessor-controlled turn-on and turn-off sequences that

may be required in certain applications.

Synchronization

In addition to sensing incoming current for OV, UV and

remote ON/OFF, the LINE-SENSE pin also monitors its pin

voltage through a 1 V threshold comparator. A pin voltage

below 1 V turns on DPA-Switch. When the voltage at LINE-

SENSE pin rises beyond 1 V to disable the output, DPA-Switch

completes its current switching cycle before the output is

forced off (similar to remote ON/OFF operation). The internal

oscillator is stopped at the end of the current cycle awaiting the

LINE-SENSE pin voltage to go low to start the next cycle. This

allows the use of the 1 V threshold to synchronize DPA-Switch

to an external signal with a frequency lower than its internal

switching frequency. A transistor or an optocoupler output

connected between the LINE-SENSE pin and the SOURCE

pin implements this function (see Figure 24). Please refer to

Figure 6 for the timing waveforms of synchronization

operation.

In order to be recognized as a synchronization pulse, the

LINE-SENSE pin needs to stay low (on-time) for at least

120 ns but no more than 2250 ns for 400 kHz (or 3080 ns for

300 kHz) internal switching frequency. In addition, the off-

time must be kept below 7.7 µs, which is a limitation set by

the lowest synchronization frequency of 128 kHz allowed by

the chip. The effective DCMAX for synchronization operation

can be calculated as 0.75 • fSYNC/fOSC. An off-time longer than

7.7 µs may force the CONTROL pin to go into the hysteretic

mode and initiate a soft-start cycle at next turn-on.

Soft-Start

Two on-chip soft-start functions are activated at start-up with

a duration of 5 ms (typical). Maximum duty cycle starts from

0% and linearly increases to the default maximum of 75% at

the end of the 5 ms duration and the current limit starts from

about 85% and linearly increases to 100% at the end of the

5 ms duration. In addition to start-up, soft-start is also activated

at each restart attempt during auto-restart and when restarting

after being in hysteretic regulation of CONTROL pin voltage

(VC), due to remote off or thermal shutdown conditions. This

effectively minimizes current and voltage stresses on the output

MOSFET, the clamp circuit and the output rectiﬁer during

start-up. This feature also helps minimize output overshoot and

prevents saturation of the transformer during start-up.

Shutdown/Auto-Restart

To minimize DPA-Switch power dissipation under fault

conditions, the shutdown/auto-restart circuit turns the power

supply on and off at an auto-restart duty cycle of typically 4%

if an out of regulation condition persists. Loss of regulation

interrupts the external current into the CONTROL pin. VC

regulation changes from shunt mode to the hysteretic auto-

restart mode as described in CONTROL pin operation section.

When the fault condition is removed, the power supply output

becomes regulated, VC regulation returns to shunt mode, and

normal operation of the power supply resumes.

Hysteretic Over-Temperature Protection

Over temperature protection is provided by a precision analog

circuit that turns the output MOSFET off when the junction

temperature exceeds the thermal shutdown temperature

(137 °C typical). When the junction temperature cools to below

the hysteretic temperature (110 °C typical), normal operation

resumes providing automatic recovery. VC is regulated in

hysteretic mode and a 4.8 V to 5.8 V (typical) sawtooth waveform

is present on the CONTROL pin while in thermal shutdown.

Bandgap Reference

All critical DPA-Switch internal voltages are derived from a

temperature-compensated bandgap reference. This reference

is also used to generate a temperature-compensated current

reference that is trimmed to accurately set the switching

frequency, current limit, and the line OV/UV thresholds.

DPA-Switch has improved circuitry to maintain all of the above

critical parameters within very tight absolute and temperature

tolerances.

High-Voltage Bias Current Source

This current source biases DPA-Switch from the DRAIN pin

and charges the CONTROL pin external capacitance during

start-up or hysteretic operation. Hysteretic operation occurs

during auto-restart, remote off and over-temperature shutdown.

In this mode of operation, the current source is switched on and

off with an effective duty cycle of approximately 20%. This

duty cycle is determined by the ratio of CONTROL pin charge

(IC(CH)) and discharge currents (ICD1 and ICD2). This current source

is turned off during normal operation when the output MOSFET

is switching. The effect of the current source switching may be

seen on the DRAIN voltage waveform as small disturbances,

which is normal.

DPA423-426

12/04 9

Using Feature Pins

FREQUENCY (F) Pin Operation

The FREQUENCY pin is a digital input pin. Shorting the

FREQUENCY pin to SOURCE pin selects the nominal

switching frequency of 400 kHz (Figure 9) which is suited for

most applications. For other applications that may beneﬁt from

lower switching frequency, a 300 kHz switching frequency can

be selected by shorting the FREQUENCY pin to the CONTROL

pin (Figure 10). This pin should not be left open.

LINE-SENSE (L) Pin Operation

When current is fed into the LINE-SENSE pin, it works as a

voltage source of approximately 2.6 V up to a maximum current

of +240 µA (typical). At +240 µA, this pin turns into a constant

current sink. Refer to Figure 8. In addition, a comparator with

a threshold of 1 V is connected at the pin and is used to detect

when the pin is shorted to the SOURCE pin.

There are a total of ﬁve functions available through the use of

the LINE-SENSE pin: OV, UV, line feed-forward with DCMAX

reduction, remote ON/OFF and synchronization. Shorting the

LINE-SENSE pin to the SOURCE pin disables all ﬁve functions.

The LINE-SENSE pin is typically used for line sensing by

connecting a resistor from this pin to the positive input DC

voltage bus to implement OV, UV and line feed-forward with

DCMAX reduction over line voltage. In this mode, the value of

the resistor determines the line OV/UV thresholds, and the

DCMAX is reduced linearly with input DC high voltage starting

from just above the UV threshold. This pin can also be used

as the input pin for remote ON/OFF and synchronization.

An external transistor placed between the LINE-SENSE pin

and the CONTROL pin realizes remote on/off via UV or OV

threshold. Synchronization is available by connecting an open

drain external MOSFET between the LINE-SENSE pin and

the SOURCE pin to generate synchronization pulse. Each

time the MOSFET turns on, the falling edge of the LINE-

SENSE pin voltage initiates a new switching cycle. The lowest

synchronization frequency guaranteed by DPA-Switch is

128 kHz. Refer to Table 2 for possible combinations of the

functions with example circuits shown in Figure 11 through

Figure 24. A description of speciﬁc functions in terms of the

LINE-SENSE pin I/V characteristic is shown in Figure 7 (right

hand side). The horizontal axis represents LINE-SENSE pin

current with positive polarity indicating currents ﬂowing into

the pin. The meaning of the vertical axes varies with functions.

For those that control the on/off states of the output such as

UV, OV and remote ON/OFF, the vertical axis represents

the enable/disable states of the output. UV triggers at IUV

(+50 µA typical with 4 µA hysteresis) and OV triggers at IOV

(+135 µA typical with 4 µA hysteresis). Between the UV and

OV thresholds, the output is enabled. For line feed-forward with

DCMAX reduction, the vertical axis represents the magnitude

of the DCMAX Line feed-forward with DCMAX reduction lowers

maximum duty cycle from 75% at IL(DC) (+55 µA typical) to

33% at IOV (+135 µA).

EXTERNAL CURRENT LIMIT (X) Pin Operation

When current is drawn out of the EXTERNAL CURRENT

LIMIT pin, it works as a voltage source of approximately

1.3 V up to a maximum current of -230 µA (typical). At

-230 µA, it turns into a constant current source (refer to Figure 8).

There are two functions available through the use of the

EXTERNAL CURRENT LIMIT pin: external current limit

and remote ON/OFF. Shorting the EXTERNAL CURRENT

LIMIT pin and SOURCE pin disables both functions. In high

efﬁciency applications, this pin can be used to reduce the current

limit externally to a value close to the operating peak current,

by connecting the pin to the SOURCE pin through a resistor.

LINE-SENSE AND EXTERNAL CURRENT LIMIT PIN TABLE*

Figure Number 11 12 13 14 15 16 17 18 19 20 21 22 23 24

Three Terminal Operation ✓

Under-Voltage ✓ ✓ ✓ ✓ ✓

Overvoltage ✓ ✓ ✓ ✓ ✓

Line Feed-Forward (DCMAX)✓ ✓ ✓ ✓

Overload Power Limiting ✓

External Current Limit ✓ ✓ ✓ ✓ ✓ ✓

Remote ON/OFF ✓ ✓ ✓ ✓ ✓ ✓

Synchronization ✓

*This table is only a partial list of many LINE-SENSE and EXTERNAL CURRENT LIMIT pin conﬁgurations that are possible.

Table 2. Typical LINE-SENSE and EXTERNAL CURRENT LIMIT Pin Conﬁgurations.

▼

DPA423-426

12/04

Figure 7. LINE-SENSE and EXTERNAL CURRENT LIMIT Pin Characteristics.

-250 -200 -150 -100 -50 50 100 150 200 2500 0

PI-2778-080801

Output

MOSFET

Switching

(Enabled)

(Disabled)

(Enabled)

(Disabled)

ILIMIT (Default)

Current

Limit

VBG

-21.5 µA

-25.5 µA

VBG + VTP

IUV(U)

IREM(U) IOV(U)

X Pin Voltage

Output

MOSFET

Switching

Maximum

Duty Cycle

L Pin Voltage

DCMAX (75%)

Note: These figures provide idealized functional characteristics with typical performance values. Please refer to the

parametric table and typical performance characteristics sections of the data sheet for measured data.

X Pin Current (µA) L Pin Current (µA)

131 µA

135 µA

47 µA

The pin can also be used as a remote on/off control input.

Table 2 shows several different ways of using this pin. See

Figure 7 for a description of the functions where the horizontal

axis (left hand side) represents the EXTERNAL CURRENT

LIMIT pin current. The meaning of the vertical axes varies

with function. For those that control the on/off states of the

output such as remote ON/OFF, the vertical axis represents the

enable/disable states of the output. For external current limit,

the vertical axis represents the magnitude of the ILIMIT

. Please

see graphs in the Typical Performance Characteristics section

for the current limit programming range and the selection of

the appropriate resistor value.

DPA423-426

12/04 11

Figure 8. LINE-SENSE (L), and EXTERNAL CURRENT LIMIT (X) Pin Input Simpliﬁed Schematic.

VBG + VT

1 V

VBG

230 µA

240 µA

CONTROL (C)

(Voltage Sense)

(Positive Current Sense - Under-Voltage,

Overvoltage, ON/OFF Maximum Duty

Cycle Reduction)

(Negative Current Sense - ON/OFF,

Current Limit Adjustment)

PI-2765-061704

DPA-Switch

LINE-SENSE (L)

EXTERNAL CURRENT LIMIT (X)

DPA423-426

12/04

Figure 9. 400 kHz Frequency Operation. Figure 10. 300 kHz Frequency Operation.

Typical Uses of FREQUENCY (F) Pin

PI-2654-071700

Input

Voltage

CONTROL

PI-2655-071700

Input

Voltage

CONTROL

Typical Uses of LINE-SENSE (L) and EXTERNAL CURRENT LIMIT (X) Pins

X F

PI-2766-070901

Input

Voltage

CONTROL

C S D

C L X S F D

Figure 11. Three Terminal Operation (LINE-SENSE and

EXTERNAL CURRENT LIMIT Features Disabled.

FREQUENCY Pin can be tied to SOURCE or

CONTROL Pin).

Figure 12. Line-Sensing for Under-Voltage, Overvoltage and

Line Feed-forward.

Figure 13. Line-Sensing for Under-Voltage Only (Overvoltage

Disabled).

Figure 14. Line-Sensing for Overvoltage Only (Under-Voltage

Disabled). Maximum Duty Cycle will be reduced at

Low Line.

PI-2767-091302

Input

Voltage

CONTROL

RLS

619 kΩ

VUV = IUV x RLS + VL (IL = IUV)

VOV = IOV x RLS + VL (IL = IOV)

For RLS = 619 kΩ

VUV = 33.3 V

VOV = 86.0 V

PI-2852-121504

Input

Voltage

D L

VUV = RLS x IUV +

VL (IL = IUV)

For Values Shown

VUV = 33.1 V

RLS

15 V

464 kΩ

150 kΩ

CONTROL

PI-2853-091302

Input

Voltage

CONTROL

RLS

1N4148

VOV = IOV x RLS +

VL (IL = IOV)

For Values Shown

VOV = 86.2 V

590 kΩ

30 kΩ

DPA423-426

12/04 13

Figure 15. Externally Set Current Limit. Figure 16. Current Limit Reduction with Line Voltage.

PI-2836-011904

Input

Voltage

RIL

For RIL = 12 kΩ

ILIMIT = 64%

See Figure 34 for other

resistor values (RIL)

For RIL = 25 kΩ

ILIMIT = 34%

CONTROL

PI-2854-050602

Input

Voltage

363 kΩ

RLS

4.2 kΩ

RIL

100% @ 36 VDC

64% @ 72 VDC

ILIMIT =

CONTROL

Figure 17. Active-on (Fail Safe) Remote ON/OFF. Figure 18. Active-off Remote ON/OFF. Maximum Duty Cycle will

be reduced.

Figure 19. Active-on Remote ON/OFF with Externally Set Current

Limit.

Figure 20. Active-off Remote ON/OFF with Externally Set Current

Limit.

ON/OFF

47 kΩ

PI-2856-072602

Input

Voltage

RIL QR

For RIL = 12 kΩ

ILIMIT = 64%

For RIL = 25 kΩ

ILIMIT = 34%

QR can be an optocoupler

output or can be replaced

by a MOSFET or manual

switch.

CONTROL

PI-2857-050602

Input

Voltage

CONTROL

47 kΩ

RMC

37.4 kΩ

QR can be an

optocoupler output

or can be replaced

by a manual switch.

ON/OFF

RIL

For RIL = 12 kΩ

ILIMIT = 64%

For RIL = 25 kΩ

ILIMIT = 34%

PI-2855-050602

Input

Voltage

CONTROL

47 kΩ

RMC

37.4 kΩ

QR can be an

optocoupler output or

can be replaced

by a manual switch.

ON/OFF

PI-2625-040501

Input

Voltage

ON/OFF

47 KΩ

QR can be an optocoupler

output or can be replaced by

a manual switch.

CONTROL

Typical Uses of LINE-SENSE (L) and EXTERNAL CURRENT LIMIT (X) Pins (cont.)

DPA423-426

12/04

PI-2858-072602

Input

Voltage

CONTROL

47 kΩ

619 kΩ

RLS

ON/OFF

For RLS = 619 kΩ

VUV = 33.3 V

VOV = 86.0 V

QR can be an optocoupler

output or can be replaced

by a MOSFET or manual

switch.

ON/OFF

47 kΩ

PI-2859-050602

Input

Voltage

CONTROL

RIL

RLS

619 kΩ

DCMAX@36 V = 75%

DCMAX@72 V = 42%

For RIL = 12 kΩ

ILIMIT = 64%

QR can be an optocoupler

output or can be replaced

by a manual switch.

Figure 21. Active-on Remote ON/OFF with LINE-SENSE and

EXTERNAL CURRENT LIMIT.

Figure 22. Active-off Remote ON/OFF with LINE-SENSE.

Figure 23. Line-Sensing and Externally Set Current Limit.

PI-3868-050602

Input

Voltage

CONTROL

ON/OFF

47 kΩ

QR can be an optocoupler

output.

For timing requirements,

see Figure 6.

Figure 24. Synchronization.

PI-2837-011904

Input

Voltage

CONTROL

RIL

RLS

12 kΩ

619 KΩ

For RLS = 619 kΩ

DCMAX@36 V = 75%

DCMAX@72 V = 42%

For RIL = 12 kΩ

ILIMIT = 64%

See Figure 34 for other

resistor values (RIL)

to select different ILIMIT

values

VUV = 33.3 V

VOV = 86.0 V

Typical Uses of LINE-SENSE (L) and EXTERNAL CURRENT LIMIT (X) Pins (cont.)

DPA423-426

12/04 15

Application Examples

Figure 25. A High Efﬁciency 30 W, 5 V, Telecom Input DC-DC Converter.

DPA425R

BAV

19WS

220 nF

VR1

SMBJ

150 C6

68 µF

10 V

1 nF

1.5 kV R14

10 Ω

18.2 kΩ

619 kΩ

1.0 Ω

C10

100 µF

10 V

C11

100 µF

10 V

C12

1 µF

10 V

36-75 VDC

1 µH

2.5 A L2

C1, C2 & C3

1 µF

100 V

D L

S X F

CONTROL

PC357N1T

PI-3472-061704

DPA-Switch

5 V, 6 A

RTN

Si4888

C9* R5*

*Optional components

R15

10 ΩR16

10 kΩ

BAV19WS

R17

10 Ω

Si4888

4.7 µF

20 V

BAV19WS

LM431AIM3

220 Ω

R11

10.0 kΩ

R10

10.0 kΩ

R12

5.1 Ω

10 kΩ

C14

1 µF

C13

10 µF

10 V

C16

100 nF

150 Ω

VIN

C17

3300 pF

High Efﬁciency 30 W Forward Converter

The circuit shown in Figure 25 is a typical implementation of a

single output DC-DC converter using DPA-Switch in a forward

conﬁguration with synchronous rectiﬁcation. This design

delivers 30 W at 5 V, from a 36 VDC to 75 VDC input with a

nominal efﬁciency at 48 VDC of 90% using the DPA425R.

By taking advantage of many of the built-in features of the

DPA-Switch, the design is greatly simpliﬁed compared to a

discrete implementation. Resistor R1 programs the input under-

voltage and overvoltage thresholds to typically 33 V and 86 V

respectively. This resistor also linearly reduces the maximum

duty from the internal maximum of 75% at 36 V to 42% at

72 V to prevent core saturation during load transients at high input

voltages. The DPA-Switch internal thresholds are toleranced

and characterized so the designer can guarantee the converter

will begin operation at 36 V, necessary to meet ETSI standards,

without the cost of an external reference IC.

The current limit is externally set by resistor R3 to just above the

drain current level needed for maximum load regulation to limit

the maximum overload power of the converter. The externally

programmable current limit feature also allows a larger

DPA-Switch family member to be selected. Using the X pin, the

current limit can be adjusted to the same level. A large device

reduces conduction losses and improves efﬁciency without

requiring any other circuit changes. This has been used here

to replace the DPA424R with a DPA425R.

The selectable 300/400 kHz switching frequency is set to 300 kHz

by connecting the FREQUENCY (F) pin to CONTROL (C).

DRAIN voltage clamping is provided by VR1, which keeps

the peak DRAIN voltage within acceptable limits. Transformer

core reset is provided by the gate capacitance of Q1 with R17

in series. Optional reset capacitance C9 with R5 can be added

if necessary to supplement the gate capacitance of Q1.

The output of the transformer is rectiﬁed using MOSFETs

to provide synchronous rectiﬁcation. The UV/OV function,

together with the turns ratio of the transformer, deﬁnes the

maximum MOSFET gate voltage, allowing the very simple

gate drive arrangement, without the need for drive windings

or a drive IC. During primary on-time, capacitor C17 couples

charge through resistor R15 to drive the gate of the forward

MOSFET, Q2. Capacitor C17 provides a DC isolated drive for

Q2, preventing gate overstress on Q1 during power down. The

time constant formed by R16 and C17 is selected to be much

longer than one switching cycle. Diode D4 resets the voltage

on capacitor C17 before the next switching cycle. During the

primary off-time, the diode D2 provides a conduction path for

the energy in inductor L2 while Q1 is still off. The transformer

DPA423-426

12/04

reset voltage on the secondary winding directly drives a positive

voltage on the gate of catch MOSFET, Q1. MOSFET Q1

provides a low loss conduction path for a substantial portion

of the primary off-time. An isolated auxiliary winding on L2,

rectiﬁed and ﬁltered by D1 and C4, provides the bias supply

for the optocoupler transistor.Output regulation is achieved

by using secondary side voltage reference, U3. The resistor

divider formed by R10 and R11, together with the reference

voltage, determines the output voltage. Diode D3 and C13 form

a soft-ﬁnish network that, together with the internal duty cycle

and current limit soft-start of the DPA-Switch, prevent output

overshoot at start-up. Resistor R7 ensures that the soft-ﬁnish

capacitor is discharged quickly when the output falls out of

regulation.Control loop response is shaped by R6, C16, R12,

C14, R9, R4 and C5, providing a wide bandwidth and good

phase margin at gain crossover. Since the PWM control in

DPA-Switch is voltage mode, no slope compensation is required

for duty cycles above 50%.

Cost Effective 6.6 W Flyback Conveter

The DPA-Switch ﬂyback power supply provides a cost effective

solution for high density PoE and VoIP DC-DC applications.

Figure 26 shows a typical implementation of a single output

ﬂyback converter using the DPA423G. For applications that

require input to output isolation, this simple, low component

count design delivers 6.6 W at 3.3 V from a 36 VDC to 57 VDC

input with a nominal efﬁciency at 48 VDC of 80%.

Resistor R2 programs the input under-voltage and overvoltage

thresholds to 33 V and 86 V respectively. Resistors R1 and R3

program the internal device current limit. The addition of line

sense resistor R1 reduces the current limit with increasing input

voltage, preventing excessive overload output current. In this

design the overload output current varies less than ±2.5% across

the entire input voltage range. Controlling the current limit also

reduces secondary component stress and leakage inductance

spikes, allowing the use of a lower VRRM (30 V rather than

40 V) Schottky output diode, D2.

The primary side Zener clamp VR1 ensures the peak drain

voltage is kept below the 220 V BVDSS rating of U1 under

input surge and overvoltage events. During normal operation,

VR1 does not conduct and C2 is sufﬁcient to limit the peak

drain voltage.

The primary bias winding provides CONTROL pin current after

start-up. Diode D3 rectiﬁes the bias winding, while components

R5 and C8 reduce high frequency switching noise and prevent

peak charging of the bias voltage. Capacitor C3 provides

local decoupling of U1 and should be physically close to the

CONTROL and SOURCE pins. Energy storage for start-up

and auto-restart timing is provided by C4.

The secondary is rectiﬁed by D2 and the Low ESR tantalum

output capacitors, C5-C7, minimizing switching ripple and

maximizing efﬁciency. A small footprint secondary output choke

L1 and ceramic output capacitor C9 are adequate to reduce high

frequency noise and ripple to below 35 mV peak-peak under

full load conditions.

The output voltage is sensed by the voltage divider formed

by resistors R8 and R9 and is fed to the low voltage 1.24 V

reference U3. Feedback compensation is provided by R6, R7

CONTROL

DPA-Switch

22 µF

10 V

0.1 µF

50 V C10

0.33 µF

C11

0.1 µF

1 µF

50 V

1 µF

10 V

330 µF

6 V

1 µH, 2A

330 µF

6 V

330 µF

6 V

1 µF

100 V

47 pF

200 V

VR1

SMAJ

150A

1 MΩ

8.66 kΩ

5.1 Ω

20 kΩ

34 kΩ

1 kΩ

100 Ω

51 Ω

619 kΩ

9, 101

6, 7

DPA423G U2

PC357

CAT431L,

SOT23

BAV19, SOD323

SL43

+VIN

36 - 57 VDC

J1-1

3.3 V, 2 A

J2-1

-VIN

J1-2

RTN

J2-2

PI-3806-061704

Figure 26. A Cost Effective 6.6 W, 3.3 V Flyback DC-DC Converter.

DPA423-426

12/04 17

and C11 together with C4 and R4. Capacitor C10 provides a

soft-ﬁnish characteristic, preventing output overshoot during

start-up.

Key Application Considerations

DPA-Switch Design Considerations

Power Table

This section provides a description of the assumptions used

to generate the power tables (Tables 1 and 3 through 6) and

explains how to use the information provided by them.

All Power tables: Tables 1 and 3 through 6

• Maximum output power is limited by the device internal

current limit. This is the peak output power which could

become the continuous output power, provided adequate

heat sinking is used.

• Data assumes adequate heat sinking to keep the junction

temperature at or below 100 °C and worst case RDS(ON) at

TJ = 100 °C.

• The use of P and G packages are recommended for device

dissipation equal to or less than 1.5 W only due to package

thermal limitation. For device dissipation above 1.5 W, use

R package.

Forward power tables: Tables 1 (upper half), 3 and 4

• Output power ﬁgures are based on forward topology using

Schottky diode rectiﬁcation. Up to 5% higher output power

is possible using synchronous rectiﬁcation.

• Dissipation data assumes a diode loss representing 6%

of the total output power and combined loss in magnetic

components representing 6% of the total output power.

DPA-Switch losses are based on a ratio between conduction

and switching losses of approximately 3:1. These

assumptions are typical for a single 5 V output forward

converter design using Schottky rectiﬁcation and adequately

designed magnetic components.

Flyback power tables: Tables 1 (lower half), 5 and 6

• Output power and dissipation ﬁgures are based on a 5 V

output using Schottky diode rectiﬁcation with an efﬁciency

of 85%. Values are generated by calculation based on

I2 • RDS(ON) losses and characterization of switching losses,

correlated to bench measurement of each DPA-Switch

device.

• Device dissipations above 1.5 W are possible using the

R package. However the forward converter topology is

recommended for such higher power designs.

The power tables provide two types of information. The ﬁrst is

the expected device dissipation for a given output power. The

second is the maximum power output. Each table speciﬁes

the input voltage range and assumes a single 5 V output using

Schottky diode rectiﬁcation.

Table 4. Output Power Table for 24 VDC Minimum Input Voltage

(See Table 3 for Notes).

Table 3. Output Power Table for 16 VDC Minimum Input Voltage.

Notes: 1. Maximum output power is limited by device internal

current limit. 2. See text in this section for a complete description of

assumptions. 3. See Part Ordering Information.

Table 5. Flyback Output Power Table for 16 VDC Minimum Input

Voltage (See Table 3 for Notes).

OUTPUT POWER TABLE

16-32 VDC RANGE (FORWARD)2

Total Device

Disipation

PRODUCT3

0.5 W 1 W 2.5 W 4 W 6 W

Max

Power

Output1

DPA423 5 W 7 W - - - 7.5 W

DPA424 7 W 10 W 15 W - - 15.5 W

DPA425 10 W 14 W 22 W 27 W 31 W

DPA426 12 W 16.5 W 25 W 31 W 37 W 43 W

OUTPUT POWER TABLE

24-48 VDC RANGE (FORWARD)2

Total Device

Disipation

PRODUCT3

0.5 W 1 W 2.5 W 4 W 6 W

Max

Power

Output1

DPA423 8 W 11 W - - - 11.5 W

DPA424 11 W 16 W 23.5 W - - 25 W

DPA425 16 W 22 W 35 W 43 W 47 W

DPA426 18 W 25 W 39 W 48 W 58 W 65 W

OUTPUT POWER TABLE

16-32 VDC RANGE (FLYBACK)2

Total Device

Disipation

PRODUCT3

0.5 W 0.75 W 1 W 1.5 W

Max

Power

Output1

DPA423 5 W - - - 6 W

DPA424 6.5 W 8.5 W 10 W - 11 W

DPA425 7 W 10 W 12 W 15 W 22 W

Table 6. Flyback Output Power Table for 24 VDC Minimum Input

Voltage.

OUTPUT POWER TABLE

24-48 VDC RANGE (FLYBACK)2

Total Device

Disipation

PRODUCT3

0.5 W 0.75 W 1 W 1.5 W

Max

Power

Output1

DPA423 7 W - - - 8.5 W

DPA424 8.5 W 11.5 W 14 W - 17 W

DPA425 -4-4-4- 34 W

DPA423-426

12/04

For example, referring to Table 1, for 36 VDC to 75 VDC input

range, DPA424 would typically dissipate 1 W in a 23 W forward

converter and has a maximum power capacity of 35 W. In the

same converter, DPA425 would dissipate 0.5 W. Selecting

DPA425 with associated reduced dissipation would increase

overall converter efﬁciency by approximately 2%.

Issues Affecting Dissipation:

1) Using synchronous rectiﬁcation will tend to reduce device

dissipation.

2) Designs with lower output voltages and higher currents

will tend to increase the device dissipation listed in the

power table.

3) Reduced input voltage decreases the available output

power for the same device dissipation. Tables 3 to 6 are

the power tables for 16 VDC and 24 VDC input voltages.

Input voltages below 16 V are possible, but since the internal

start-up current source is not speciﬁed at voltages below 16

V, an external chip supply current should be fed into the

CONTROL pin approximately equal to but less than ICD1.

DPA-Switch Selection

Use Tables 1 and 3 through 6 to select the DPA-Switch based

on device dissipation. Selecting the optimum DPA-Switch

depends upon required maximum output power, efﬁciency, heat

sinking constraints and cost goals. With the option to externally

reduce current limit, a larger DPA-Switch may be used for

lower power applications where higher efﬁciency is needed

or minimal heat sinking is available. Generally, selecting the

next larger device, than is required for power delivery will give

the highest efﬁciency. Selecting even larger devices may give

little or no improvement in efﬁciency due to the improvement

in conduction losses being negated by larger device switching

losses. Figure 50 provides information on switching losses.

This together with conduction loss calculations give an estimate

of device dissipation.

Primary Clamp

A primary clamp network is recommended to keep the peak

DRAIN voltage due to primary leakage inductance to below

the BVDSS speciﬁcation. A Zener diode combined with a small

value capacitor connected across the primary winding is a low

cost and low part count implementation.

Output Rectiﬁcation

Rectiﬁcation of the secondary is typically performed using

Schottky diodes or synchronous rectiﬁcation. Schottky diodes

are selected for peak inverse voltage, output current, forward drop

and thermal conditions. Synchronous rectiﬁcation requires the

additional complication of providing gate drive. The speciﬁed

line under-voltage and line overvoltage thresholds of DPA-Switch

simpliﬁes deriving gate drive directly from the transformer

secondary winding for many applications. The turns ratio of

the transformer together with the under/over-voltage thresholds

deﬁnes the minimum and maximum gate voltages, removing

the need for Zeners to clamp the gate voltage.

Soft-Start

Generally a power supply experiences maximum stress at

start-up before the feedback loop achieves regulation. For a

period of 5 ms the on-chip soft-start linearly increases the duty

cycle from zero to the default DCMAX at turn-on. In addition,

the primary current limit increases from 85% to 100% over the

same period. This causes the output voltage to rise in an orderly

manner allowing time for the feedback loop to take control of

the duty cycle. This integrated soft-start reduces the stress on

the DPA-Switch MOSFET, clamp circuit and output diode(s),

and helps prevent transformer saturation during start-up. Also,

soft-start limits the amount of output voltage overshoot, and in

many applications eliminates the need for a soft-ﬁnish capacitor.

If necessary, to remove output overshoot, a soft-ﬁnish capacitor

may be added to the secondary reference.

Switching Frequency

The FREQUENCY pin of DPA-Switch offers a switching

frequency option of 400 kHz or 300 kHz. Operating at 300 kHz

will increase the amount of magnetization energy stored in the

transformer. This is ideal for applications using synchronous

rectiﬁcation driven directly from the transformer secondary

where this energy can be used to drive the catch MOSFET

gate.

Transformer Design

It is recommended that the forward converter transformer be

designed for maximum operating ﬂux swing of 1500 Gauss

and a peak ﬂux density of 3500 Gauss. When operating at the

maximum current limit of the selected DPA-Switch (during

overload conditions), neither magnetic component (transformer

and output inductor) should be allowed to saturate. When a larger

device than necessary has been selected, reducing the internal

current limit close to the operating peak current limits overload

power and minimizes the size of the secondary components.

No-load and Standby Consumption

Cycle skipping operation at light or no load can signiﬁcantly

reduce power loss. In addition this operating mode ensures

that the output maintains regulation even without an external

minimum load. However, if cycle skipping is undesirable in

a particular application, it can be avoided by adding sufﬁcient

pre-load.

DPA-Switch Layout Considerations

The DPA-Switch can operate with large DRAIN current, the

following guidelines should be carefully followed.

Primary Side Connections

The tab of DPA-Switch R package is the intended return path

DPA423-426

12/04 19

for the high switching currents. Therefore, the tab should be

connected by wide, low impedance traces back to the input

decoupling capacitor. The SOURCE pin should not be used

to return the power currents; incorrect operation of the device

may result. The SOURCE is only intended as a signal ground.

The device tab (SOURCE) is the correct connection for power

currents with the R package.

The CONTROL pin bypass capacitor should be located as

close as possible to the SOURCE and CONTROL pins and its

SOURCE connection trace should not be shared by the main

MOSFET switching currents. All SOURCE pin referenced

components connected to the LINE-SENSE or EXTERNAL

CURRENT LIMIT pins should also be located closely between

their respective pin and SOURCE. Once again, the SOURCE

connection trace of these components should not be shared

by the main MOSFET switching currents. It is critical that

the tab (SOURCE) power switching currents are returned to the

input capacitor through a separate trace that is not

shared by the components connected to CONTROL,

LINE-SENSE or EXTERNAL CURRENT LIMIT pins.

Any traces to the L or X pins should be kept as short as possible

and away from the DRAIN trace to prevent noise coupling.

LINE-SENSE resistor (R1 in Figure 25) should be located close

to the L pin to minimize the trace length on the L pin side.

In addition to the CONTROL pin capacitor (C6 in Figure 25),

a high frequency bypass capacitor in parallel is recommended

as close as possible between SOURCE and CONTROL pins for

better noise immunity. The feedback optocoupler output should

also be located close to the CONTROL and SOURCE pins of

DPA-Switch.

Heat Sinking

To maximize heat sinking of the DPA-Switch R or G package

and the other power components, special thermally conductive

PC board material (aluminum clad PC board) is recommended.

This has an aluminum sheet bonded to the PC board during

the manufacturing process to provide heat sinking directly

and allow the attachment of an external heat sink. If normal

PC board material is used (such as FR4), providing copper

areas on both sides of the board and using thicker copper will

improve heat sinking.

If an aluminum clad board is used then shielding of switching

nodes is recommended. This consists of an area of copper placed

directly underneath switching nodes such as the drain node,

and output diode to provide an electrostatic shield to prevent

coupling to the aluminum substrate. These areas are connected

to input negative in the case of the primary and output return

for secondary. This reduces the amount of capacitive coupling

into the insulated aluminum substrate that can then appear on

the output as ripple and high frequency noise.

Quick Design Checklist

As with any power supply design, all DPA-Switch designs

should be veriﬁed on the bench to make sure that component

speciﬁcations limits are not exceeded under worst case

conditions. The following minimum set of tests for DPA-Switch

forward converters is strongly recommended:

1. Maximum drain voltage – Verify that peak VDS does not

exceed minimum BVDSS at highest input voltage and

maximum overload output power. It is normal, however, to

have additional margin of approximately 25 V below BVDSS

to allow for other power supply component unit-to-unit

variations. Maximum overload output power occurs when

the output is loaded to a level just before the power supply

goes into auto-restart (loss of regulation).

2. Transformer reset margin – Drain voltage should also be

checked at highest input voltage with a severe load step

(50-100%) to verify adequate transformer reset margin. This

test shows the duty cycle at high input voltage, placing the

most demand on the transformer reset circuit.

3. Maximum drain current – At maximum ambient temperature,

maximum input voltage and maximum output load, verify

drain current waveforms at start-up for any signs of

transformer or output inductor saturation and excessive

leading edge current spikes. DPA-Switch has a leading edge

blanking time of 100 ns to prevent premature termination

of the on cycle. Verify that the leading edge current spike

does not extend beyond the blanking period.

4. Thermal check – At maximum output power, minimum

input voltage and maximum ambient temperature, verify

that temperature speciﬁcations are not exceeded for the

transformer, output diodes, output choke(s) and output

capacitors. The DPA-Switch is fully protected against over-

temperature conditions by its thermal shutdown feature. It

is recommended that sufﬁcient heat sinking is provided to

keep the tab temperature at or below 115 °C (R package),

SOURCE pins at or below 100 °C (P/G packages) under

worst case continuous load conditions (at low input voltage,

maximum ambient and full load). This provides adequate

margin to minimum thermal shutdown temperature

(130 °C) to account for part-to-part RDS(ON) variation. When

monitoring device temperatures, note that the junction-

to-case thermal resistance should be accounted for when

estimating die temperature.

Design Tools

Up-to-date information on design tools is available at the Power

Integrations website: www.powerint.com.

DPA423-426

12/04

Figure 27. Layout Considerations for DPA-Switch Using R Package.

PI-2883-060602

Solder Side

Component Side

TOP VIEW

Transformer

Out

Maximize hatched copper area for optimum heat sinking

Via between board layers

Opto-

coupler

Inductor

(Coupled)

Output

Diode

DPA-Switch

Figure 28. Layout Considerations for DPA-Switch Using G Package.

PI-3805-012904

Opto-

coupler

Out

DPA-Switch Heatsink

Bottom Diode

Heatsink

C L X F

S D S

DPA-Switch

Top Side PCB

Bottom Side PCB

(Two sided printed circuit board)

TOP VIEW

Maximize hatched copper area for optimum heat sinking

Via between board layers

DPA423-426

12/04 21

Parameter Symbol

Conditions

SOURCE = 0 V; TJ = -40 to 125 °C

See Figure 32

(Unless Otherwise Speciﬁed)

Min Typ Max Units

CONTROL FUNCTIONS

Switching

Frequency fOSC TJ = 25 °C

FREQUENCY Pin

Connected to SOURCE 375 400 425

kHz

FREQUENCY Pin

Connected to SOURCE 282 300 317

Duty Cycle (Prior

to Cycle Skipping) DCMIN 4 6 %

Maximum Duty

Cycle DCMAX IC = ICD1

VL = 0 V 71 75 79

IL = 80 µA52 62 71

IL = 115 µA32 42 57

Control Current

at Start of Cycle

Skipping

IC(skip) TJ = 25 °C; fOSC = 400 kHz

DPA423 7.2 9.0

DPA424 8.2 10.0

DPA425 10.0 12.0

DPA426 11.5 14.0

External Bias

Current IBTJ = 25 °C; fOSC = 400 kHz

DPA423 2 2.8 3.5

DPA424 2.5 3.5 4.4

DPA425 3.6 4.8 6.0

DPA426 4.4 5.7 7.1

Softstart Time tSOFT TJ = 25 °C; DCMIN to DC MAX 5 7.2 ms

PWM Gain DCreg TJ = 25 °C; IC = (IC(skip) + IB)/2 -28 -22 -18 %/mA

ABSOLUTE MAXIMUM RATINGS(1,4)

DRAIN Voltage .................................. ................ -0.3 V to 220 V

DRAIN Peak Current: DPA423......................................1.75 A

DPA424....................................... 3.5 A

DPA425........................................... 7A

DPA426....................................... 9.6 A

CONTROL Voltage ................................................ -0.3 V to 9 V

CONTROL Current .................................................... 100 mA

LINE SENSE Pin Voltage ...................................... -0.3 V to 9 V

EXTERNAL CURRENT LIMIT Pin Voltage ........ -0.3 V to 9 V

FREQUENCY Pin Voltage .................................... -0.3 V to 9 V

Storage Temperature .......................................... -65 °C to 150 °C

Operating Junction Temperature(2) ..................... -40 °C to 150 °C

Lead Temperature(3) ...................................................... 260 °C

Notes:

1. All voltages referenced to SOURCE, TA = 25 °C.

2. Normally limited by internal circuitry.

3. 1/16” from case for 5 seconds.

4. Maximum ratings speciﬁed may be applied, one at a time,

without causing permanent damage to the product.

Exposure to Absolute Maximum Rating conditions for

extended periods of time may affect product reliability.

THERMAL IMPEDANCE

Thermal Impedance: P or G Package:

(θJA) ........................... 70 °C/W(1); 60 °C/W(2)

(θJC)(3) ............................................... 11 °C/W

R Package:

(θJA) ............................................... 40 °C/W(4)

............................................... 30 °C/W(5)

(θJC)(6) .................................................. 2 °C/W

Notes:

1. Soldered to 0.36 sq. in. (232 mm2), 2 oz. (610 g/m2) copper clad.

2. Soldered to 1 sq. in. (645 mm2), 2 oz. (610 g/m2) copper clad.

3. Measured on pin 7 (SOURCE) close to plastic interface.

4. Soldered to 1 sq. in. (645 mm2), 2 oz. (610 g/m2) copper clad.

5. Soldered to 3 sq. in. (1935 mm2), 2 oz. (610 g/m2) copper clad.

6. Measured at the back surface of tab.

DPA423-426

12/04

Parameter Symbol

Conditions

SOURCE = 0 V; TJ = -40 to 125 °C

See Figure 32

(Unless Otherwise Speciﬁed)

Min Typ Max Units

CONTROL FUNCTIONS

PWM Gain

Temperature Drift See Note A -0.01 %/mA/°C

Dynamic

Impedance ZCTJ = 25 °C; IC = (IC(skip) + IB)/2 10 15 22 Ω

Dynamic

Impedance

Temperature Drift

0.18 %/°C

CONTROL Pin

Internal Filter Pole 30 kHz

SHUTDOWN/AUTO-RESTART

CONTROL Pin

Charging Current IC(CH)

During Startup and Auto-Restart:

VC = 5.0 V; VD = 16 V & 40 V; TJ = 25 °C -5.2 -4 -3

Average Current at the Beginning of

Softstart: VC = 5.0 V;

VD = 16 V & 40 V; TJ = 25 °C

-19

Charging Current

Temperature Drift See Note A -0.6 %/°C

Auto-Restart

Upper Threshold

Voltage

VC(AR)U 5.8 V

Auto-Restart

Lower Threshold

Voltage

VC(AR)L 4.5 4.8 5.1 V

Auto-Restart

Hysteresis Voltage VC(AR)Hyst 0.8 1 V

Auto-Restart Duty

Cycle DC(AR)

CCONTROL = 22 µF; fOSC = 400 kHz;

VX = 0 V 10 %

Auto-Restart

Frequency f(AR)

CCONTROL = 22 µF; fOSC = 400 kHz;

VX = 0 V 3.8 Hz

LINE-SENSE (L) AND EXTERNAL CURRENT LIMIT (X) INPUTS

Line Under-

Voltage Threshold

Current and

Hysteresis (L Pin)

IUV TJ = 25 °C

Threshold from Off to On 48 50 52

µAThreshold from On to Off 44.5 47 49.5

Hysteresis 2 3

DPA423-426

12/04 23

Parameter Symbol

Conditions

SOURCE = 0 V; TJ = -40 to 125 °C

See Figure 32

(Unless Otherwise Speciﬁed)

Min Typ Max Units

LINE-SENSE (L) AND EXTERNAL CURRENT LIMIT (X) INPUTS (cont.)

Line Overvoltage

or Remote ON/

OFF Threshold

Current and

Hysteresis (L Pin)

IOV TJ = 25 °C

Threshold from On to Off 135 149

µAThreshold from Off to On 117 131

Hysteresis 4

Remote ON/OFF

Negative Thresh-

old Current and

Hysteresis

(X Pin)

IREM TJ = 25 °C

Threshold from On to Off -27 -21.5 -16

µAThreshold from Off to On -25.5

Hysteresis 4.5

L Pin Short Circuit

Current IL(SC)

VL = VC175 240 380 µA

VL = 0 V -230 -170

X Pin Short Circuit

Current IX(SC) VX = 0 V Normal Mode -270 -230 -185 µA

Remote OFF using L Pin -105 -85 -65

Line Pin Voltage

(Positive Current) VL

IL = IUV 2.05 2.35 2.6

IL = IOV 2.1 2.5 2.9

X Pin Voltage

(Negative Current) VX

IX = -50 µA 1.35

IX = -150 µA 1.25

Maximum Duty

Cycle Reduction

Onset Threshold

Current

IL(DC) TJ = 25 °C 55 µA

Remote OFF

DRAIN Supply

Current

ID(RMT)

VD = 40 V

VX = 0 V

VL = Floating 0.6 1.1

VL = VC0.9 1.5

L Pin Voltage

Turn-On Thresh-

old in Synchro-

nous Mode

VL(TH) 0.6 1 1.4 V

On-Time Pulse

Width for

Synchronization

ton(sync)

fOSC = 400 kHz 120 2250

fOSC = 300 kHz 120 3080

Off-Time Pulse

Width for

Synchronization

toff(sync) 0.25 7.7 µs

Synchronous

Turn On Delay tdelay(sync)

From Synchronous On to Drain

Turn-On 250 ns

DPA423-426

12/04

Parameter Symbol

Conditions

SOURCE = 0 V; TJ = -40 to 125 °C

See Figure 32

(Unless Otherwise Speciﬁed)

Min Typ Max Units

FREQUENCY (F) INPUT

FREQUENCY Pin

Threshold Voltage VF1.1 4 V

FREQUENCY Pin

Input Current IF

VF = 0 V -0.38 µA

VF = VC17 120

FREQUENCY Pin

Delay Time tdelay(VF) 2µs

CIRCUIT PROTECTION

Self Protection

Current Limit (See

Note B)

ILIMIT TJ = 25 °C

DPA423 di/dt = 300 mA/µs 1.16 1.25 1.34

DPA424 di/dt = 600 mA/µs 2.32 2.50 2.68

DPA425 di/dt = 1.25 A/µs 4.65 5.00 5.35

DPA426 di/dt = 1.75 A/µs 6.50 7.00 7.50

Initial Current

Limit IINIT VD = 35 V 0.9 x

ILIMIT(min)

Leading Edge

Blanking Time tLEB TJ = 25 °C 100 ns

Current Limit

Delay tIL(D) IC = (IC)skip) + IB)/2 100 ns

Thermal Shut-

down Temperature TJ(SD) 130 137 145 °C

Thermal Shut-

down Hysteresis TJ(SD)hyst 27 °C

Power-Up Reset

Threshold Voltage VC(RESET) 1.5 2.75 4 V

OUTPUT

ON-State

Resistance RDS(ON)

DPA423

ID = 300 mA

TJ = 25 °C 1.30 1.50

Ω

TJ = 100 °C 2.00 2.30

DPA424

ID = 600 mA

TJ = 25 °C 0.65 0.75

TJ = 100 °C 1.00 1.15

DPA425

ID = 1.25 A

TJ = 25 °C 0.33 0.38

TJ = 100 °C 0.50 0.58

DPA426

ID = 1.75 A

TJ = 25 °C 0.24 0.28

TJ = 100 °C 0.37 0.43

OFF-State Drain

Leakage Current IDSS

VX, VL = Floating;

VD = 150 V;

TJ = 125 °C;

IC = (IC(skip) + IB)/2

DPA423 65

µA

DPA424 130

DPA425 260

DPA426 360

DPA423-426

12/04 25

Parameter Symbol

Conditions

SOURCE = 0 V; TJ = -40 to 125 °C

See Figure 32

(Unless Otherwise Speciﬁed)

Min Typ Max Units

OUTPUT (cont.)

Breakdown

Voltage BVDSS

VX, VL = Floating; TJ = 25 °C;

IC = (IC(skip) + IB)/2; See Note C 220 V

Rise Time tRMeasured in a Typical Application 10 ns

Fall Time tFMeasured in a Typical Application 10 ns

SUPPLY VOLTAGE CHARACTERISTICS

DRAIN Supply

Voltage See Note D 16 V

Shunt Regulator

Voltage VC(SHUNT) IC = (IC(skip) + IB)/2: TJ = 25 °C 5.6 5.85 6.0 V

Shunt Regulator

Temperature Drift IC = (IC(skip) + IB)/2 ±50 PPM/°C

CONTROL

Supply/Discharge

Current

ICD1

Output

MOSFET Enabled

VL = 0 V;

fOSC = 400 kHz

DPA423 1.9 2.3 2.7

DPA424 2.6 3.0 3.4

DPA425 3.7 4.3 4.8

DPA426 4.8 5.4 6

Output MOSFET Disabled

VL = 0 V; fOSC = 400 kHz 0.4 0.73 1.2

ICD2

NOTES:

A. For speciﬁcations with negative values, a negative temperature coefﬁcient corresponds to an increase in

magnitude with increasing temperature, and a positive temperature coefﬁcient corresponds to a decrease in

magnitude with increasing temperature.

B. For externally adjusted current limit values, please refer to Figure 34 (Current Limit vs. External Current Limit

Resistance) in the Typical Performance Characteristics section.

C. Breakdown voltage may be checked against minimum BVDSS speciﬁcation by ramping the DRAIN pin voltage up

to but not exceeding minimum BVDSS.

D. It is possible to start up and operate

DPA-Switch

at DRAIN voltages well below 16 V. However, the CONTROL

pin charging current is reduced, which affects start-up time, auto-restart frequency, and auto-restart duty cycle.

Refer to Figure 44, the characteristic graph on CONTROL pin charge current (IC) vs. DRAIN voltage for low

voltage operation characteristics.

DPA423-426

12/04

PI-2039-033001

DRAIN

VOLTAGE

0 V

90%

10%

90%

D = t1

1.4

1.2

0.6

0.2

0.4

0.8

0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2

Time (µs)

DRAIN Current (Normalized)

PI-2889-061302

tLEB (Blanking Time) +

tIL(D) (Current Limit Delay)

Temperature Range:

-40 °C to 125 °C

120

100

0 2 4 6 8 10

CONTROL Pin Voltage (V)

CONTROL Pin Current (mA)

Slope

Dynamic

Impedance =

PI-2880-060302

PI-2860-050602

5-50 V

40 V

0-15 V

0.1 µF47 µF

470 Ω

5 W

470 Ω

0-200 kΩ

0-70 kΩ

NOTE: This test circuit is not applicable for current limit or output characteristic measurements.

SF X

CONTROL

DPA-Switch

Figure 29. Duty Cycle Measurement.

Figure 30. CONTROL Pin I-V Characteristic. Figure 31. Typical Drain Operation Current Envelope.

Figure 32. DPA-Switch General Test Circuit.

DPA423-426

12/04 27

The following precautions should be followed when testing

DPA-Switch by itself outside of a power supply. The schematic

shown in Figure 32 is suggested for laboratory testing of

DPA-Switch.

When the DRAIN pin supply is turned on, the part will be in

the auto-restart mode. The CONTROL pin voltage will be

oscillating at a low frequency between 4.8 V and 5.8 V and

the drain is turned on every eighth cycle of the CONTROL pin

oscillation. If the CONTROL pin power supply is turned on

while in this auto-restart mode, there is only a 12.5% chance

that the CONTROL pin oscillation will be in the correct state

(drain active state) so that the continuous drain voltage waveform

may be observed. It is recommended that the VC power supply

be turned on ﬁrst and the DRAIN pin power supply second if

continuous drain voltage waveforms are to be observed. The

12.5% chance of being in the correct state is due to the divide-

by-8 counter. Temporarily shorting the CONTROL pin to the

SOURCE pin will reset DPA-Switch, which then will come up

in the correct state.

BENCH TEST PRECAUTIONS FOR EVALUATION OF ELECTRICAL CHARACTERISTICS

Typical Performance Characteristics

0.3

0.2

0.1

0.5

0.4

0.6

0.7

0.8

0.9

1.0

1.1

0 5 10 15 20 25 30 35

External Current Limit Resistor RIL (kΩ)

Current Limit

(Normalized to Internal Current Limit)

PI-2839-032202

Minimum

Typical

Maximum

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1.1

1.0

-230 -180 -130 -80 -30

IX (µA)

Current Limit

(Normalized to Internal Current Limit)

PI-2838-032202

Figure 33. Current Limit vs. EXTERNAL CURRENT LIMIT Pin Current.

Figure 34. Current Limit vs. External Current Limit Resistance.

DPA423-426

12/04

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-2868-051502

Output Frequency

(Normalized to 25 °C)

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-2866-051702

Current Limit

(Normalized to 25 °C)

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-2872-051602

Current Limit

(Normalized to 25 °C)

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-2865-051702

Overvoltage Threshold

(Normalized to 25 °C)

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-2867-051502

Under-Voltage Threshold

(Normalized to 25 °C)

7.0

4.0

6.0

5.0

0.0

-200 -100 -50-150 0 50 100 150 200 250

LINE-SENSE Pin Current (µA)

LINE-SENSE Pin Voltage (V)

PI-2871-051502

2.0

1.0

3.0

Typical Performance Characteristics (cont.)

Figure 35. Frequency vs. Temperature. Figure 36. Internal Current Limit vs. Temperature.

Figure 37. External Current Limit vs. Temperature with

RIL = 12 kΩ.

Figure 38. Overvoltage Threshold vs. Temperarture .

Figure 39. Under-Voltage Threshold vs. Temperature. Figure 40. LINE-SENSE Pin Voltage vs. Current.

DPA423-426

12/04 29

1.6

1.0

1.4

1.2

0-225

-250 -150

-175

-200 -125 -50 -25

-100-75 0

EXTERNAL CURRENT LIMIT Pin Current (µA)

EXTERNAL CURRENT LIMIT

Pin Voltage (V)

PI-2869-051502

0.4

0.2

0.6

0.8

VX = 1.375 - IXx 843 kΩ

-200 µA ≤ IX ≤ -25 µA

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-2863-011904

CONTROL Pin Current

(Normalized to 25 °C)

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-2864-051702

Onset Threshold Current

(Normalized to 25 °C)

5.0

3.5

3.0

4.0

4.5

0.0

0 10 20 30 40 50 60

DRAIN Voltage (V)

CONTROL Pin

Charging Current (mA)

PI-2841-051502

1.0

0.5

1.5

2.5

2.0

VC = 5 V

1.2

0.8

1.0

-50 -20 0 25 50 75 100 125 150

Junction Temperature (°C)

Remote OFF DRAIN Supply Current

(Normalized to 25 °C)

PI-2870-051502

0.2

0.4

0.6

1.1

1.0

0.9

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

Breakdown Voltage

(Normalized to 25 °C)

PI-176B-033001

Typical Performance Characteristics (cont.)

Figure 41. EXTERNAL CURRENT LIMIT Pin Voltage

vs. Current

Figure 42. CONTROL Pin Current at Minimum Duty

Cycle vs. Temperature.

Figure 43. Max. Duty Cycle Reduction Onset Threshold

Current vs. Temperature.

Figure 44. IC vs. DRAIN Voltage.

Figure 45. LINE-SENSE Pin Voltage vs. Current. Figure 46. LINE-SENSE Pin Voltage vs. Current.

DPA423-426

12/04

Typical Performance Characteristics (cont.)

0 1 2 3 4 5

Drain Voltage (V)

PI-2849-050302

Drain Current (A)

Scaling Factors:

DPA423 = 0.18

DPA424 = 0.36

DPA425 = 0.72

DPA426 = 1.00

TCASE = 25 °C

TCASE = 100 °C

2.5

1.5

0.5

-50 -25 0 25 7550 100 125 150

Temperature (°C)

PI-2850-011904

On-resistance

(Normalized to 25 °C)

0 20 40 60 80 100 120 140 160

100

1000

10000

PI-2851-011904

DRAIN Voltage (V)

Capacitance (pF)

Scaling Factors:

DPA423 = 0.18

DPA424 = 0.36

DPA425 = 0.72

DPA426 = 1.00

2.5

1.5

0.5

0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1

In-Circuit Peak Drain Current Normalized to

Nominal Device Internal Current Limit

PI-2877-061704

Switching Loss (W) at 400 kHz

Scaling Factors:

DPA423 = 0.08

DPA424 = 0.20

DPA425 = 0.55

DPA426 = 1.00

Data taken from a typical

forward converter design.

Figure 47. Output Characteristics. Figure 48. On-Resistance vs. Temperature.

Figure 49. COSS vs. DRAIN Voltage. Figure 50. Typical Switching Loss.

DPA423-426

12/04 31

PART ORDERING INFORMATION

DPA-Switch Product Family

Series Number

Package Identiﬁer

G Plastic Surface Mount DIP (423, 424 & 425 only)

P Plastic DIP

R Plastic TO-263-7C (available only with TL option)

Lead Finish

Blank Standard (Sn Pb)

N Pure Matte Tin (Pb-Free)

Tape & Reel and Other Options

Blank Standard Conﬁgurations

TL Tape & Reel, (G Package: 1000 min., R Package: 750

min.)

DPA 423 G N - TL

DPA423-426

12/04

PI-2076-101102

8 5

DIP-8

D S .004 (.10)

-E-

-D-

-F-

DIM

inches

0.367-0.387

0.240-0.260

0.125-0.145

0.015-0.040

0.120-0.140

0.057-0.068

0.014-0.022

0.008-0.015

0.100 BSC

0.030 (MIN)

0.300-0.320

0.300-0.390

0.300 BSC

9.32-9.83

6.10-6.60

3.18-3.68

0.38-1.02

3.05-3.56

1.45-1.73

0.36-0.56

0.20-0.38

2.54 BSC

0.76 (MIN)

7.62-8.13

7.62-9.91

7.62 BSC

Notes:

1. Package dimensions conform to JEDEC

specification MS-001-AB for standard dual in-line

(DIP) package .300 inch row spacing

(PLASTIC) 8 leads (issue B, 7/85).

2. Controlling dimensions are inches.

3. Dimensions shown do not include mold flash

or other protrusions. Mold flash or

protrusions shall not exceed .006 (.15) on any

side.

4. D, E and F are reference datums on the molded

body.

QP08A

PI-2077-041003

8 5

G08A

SMD-8

D S .004 (.10)

E S .010 (.25)

-E-

-D-

-F-

DIM

inches

0.367-0.387

0.240-0.260

0.125-0.145

0.004-0.012

0.036-0.044

0.057-0.068

0.048-0.053

0.032-0.037

0.007-0.011

0.010-0.012

0.100 BSC

0.030 (MIN)

0.372-0.388

0-8°

9.32-9.83

6.10-6.60

3.18-3.68

0.10-0.30

0.91-1.12

1.45-1.73

1.22-1.35

0.81-0.94

0.18-0.28

0.25-0.30

2.54 BSC

0.76 (MIN)

9.45-9.86

0-8°

Notes:

1. Package dimensions conform to JEDEC

specification MS-001-AB (issue B, 7/85)

except for lead shape and size.

2. Controlling dimensions are inches.

3. Dimensions shown do not include mold

flash or other protrusions. Mold flash

or protrusions shall not exceed .006 (.15)

on any side.

4. D, E and F are reference datums on the

molded body.

.004 (.10)

.010 (.25) M A S

.420

.046 .060 .060 .046

.080

Pin 1

.086

.186

.286

Solder Pad Dimensions

DPA423-426

12/04 33

.165 (4.19)

.185 (4.70)

R07C

TO-263-7C

PI-2664-122004

-A-

LD #1

.580 (14.73)

.620 (15.75)

.390 (9.91)

.420 (10.67)

.326 (8.28)

.336 (8.53)

.055 (1.40)

.066 (1.68)

.100 (2.54)

REF

0.68 (1.73)

MIN

.208 (5.28)

Ref.

.024 (0.61)

.034 (0.86)

.225 (5.72)

MIN

.245 (6.22)

MIN

.000 (0.00)

.010 (0.25)

.090 (2.29)

.110 (2.79)

.012 (0.30)

.024 (0.61)

°8 -°0

.045 (1.14)

.055 (1.40)

.050 (1.27)

Notes:

1. Package Outline Exclusive of Mold Flash & Metal Burr.

2. Package Outline Inclusive of Plating Thickness.

3. Foot Length Measured at Intercept Point Between

Datum A Lead Surface.

4. Controlling Dimensions are in Inches. Millimeter

Dimensions are shown in Parentheses.

5. Minimum metal to metal spacing at the package body

for the omitted pin locations is .068 in. (1.73 mm).

.004 (0.10)

.315 (8.00)

.128 (3.25)

.038 (0.97)

.050 (1.27)

.380 (9.65)

.638 (16.21)

Solder Pad

Dimensions

DPA423-426

12/04

DPA423-426

12/04 35

Revision Notes Date

F 1) Final release data sheet. 6/02

G 2) Updated Figure 25 and text description. 9/02

H 1) Corrected missing text on page 9 and corrected Table 4.

2) Updated R package description.

3) Revised thermal impedances (θJA), DCMIN, VF

, IF

, and BVDSS.

4) Updated Figure 25 and description in text.

4/03

I 1) Corrected text errors on pp. 1, 7 and 20. 5/03

J 1) Figure 25 and description in text updated. 5/03

K 1) Added P and G packages. 1/04

L 1) Corrected Figure 3 4/04

M 1) Added package information to Table 1.

2) Revised Figure 13.

3) Added lead-free ordering information

12/04

DPA423-426

12/04

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described herein nor does it convey any license under its patent rights or the rights of others. POWER INTEGRATIONS MAKES NO WARRANTIES HEREIN

AND SPECIFICALLY DISCLAIMS ALL WARRANTIES INCLUDING, WITHOUT LIMITATION, THE IMPLIED WARRANTIES OF MERCHANTABILITY,

FITNESS FOR A PARTICULAR PURPOSE, AND NON-INFRINGEMENT OF THIRD PARTY RIGHTS.

PATENT INFORMATION

The products and applications illustrated herein (including circuits external to the products and transformer construction) may be covered by one or more U.S.

and foreign patents or potentially by pending U.S. and foreign patent applications assigned to Power Integrations. A complete list of Power Integrationsʼ patents

may be found at www.powerint.com.

LIFE SUPPORT POLICY

POWER INTEGRATIONSʼ PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS

WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF POWER INTEGRATIONS. As used herein:

1. Life support devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, and whose failure to perform, when

properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a signiﬁcant injury to the user.

2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life

support device or system, or to affect its safety or effectiveness.

The PI logo, TOPSwitch, TinySwitch, LinkSwitch, DPA-Switch and EcoSmart are registered trademarks of

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