LM337TLF01 - NATIONAL SEMICONDUCTOR

LM2696

LM2696 3A, Constant On Time Buck Regulator

Literature Number: SNVS375A

LM2696

May 18, 2009

3A, Constant On Time Buck Regulator

General Description

The LM2696 is a pulse width modulation (PWM) buck regu-

lator capable of delivering up to 3A into a load. The control

loop utilizes a constant on-time control scheme with input

voltage feed forward. This provides a topology that has ex-

cellent transient response without the need for compensation.

The input voltage feed forward ensures that a constant

switching frequency is maintained across the entire VIN range.

The LM2696 is capable of switching frequencies in the range

of 100 kHz to 500 kHz. Combined with an integrated 130

mΩ high side NMOS switch the LM2696 can utilize small

sized external components and provide high efficiency. An

internal soft-start and power-good flag are also provided to

allow for simple sequencing between multiple regulators.

The LM2696 is available with an adjustable output in an ex-

posed pad TSSOP-16 package.

Features

■Input voltage range of 4.5V–24V

■Constant On-Time

■No compensation needed

■Maximum Load Current of 3A

■Switching frequency of 100 kHz–500 kHz

■Constant frequency across input range

■TTL compatible shutdown thresholds

■Low standby current of 12 µA

■130 mΩ internal MOSFET switch

Applications

■High efficiency step-down switching regulators

■LCD Monitors

■Set-Top Boxes

Typical Application Circuit

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LM2696 3A, Constant On Time Buck Regulator

Connection Diagram

Top View

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eTSSOP-16 Package

Pin Descriptions

Pin # Name Function

1, 2, 3 SW Switching node

4 CBOOT Bootstrap capacitor input

5 AVIN Analog voltage input

6 EXTVCC Output of internal regulator for decoupling

7 FB Feedback signal from output

8 N/C No connect

9 GND Ground

10 SS Soft-start pin

11 PGOOD Power-good flag, open drain output

12 RON Sets the switch on-time dependent on current

13 SD Shutdown pin

14, 15, 16 PVIN Power voltage input

- Exposed Pad Must be connected to ground

Ordering Information

Order Number Package Type NSC Package

Drawing Supplied As

LM2696MXA eTSSOP-16 MXA16A 92 units/rail

LM2696MXAX eTSSOP-16 MXA16A 2,500 Units Tape and Reel

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LM2696

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

Voltages from the indicated pins to GND

AVIN −0.3V to +26V

PVIN −0.3V to (AVIN+0.3V)

CBOOT −0.3V to +33V

CBOOT to SW −0.3V to +7V

FB, SD, SS, PGOOD −0.3V to +7V

Storage Temperature Range −65°C to +150°C

Junction Temperature +150°C

Lead Temperature

(Soldering, 10 sec.) 260°C

Minimum ESD Rating 1.5 kV

Operating Range

Junction

Temperature

−40°C to +125°C

AVIN to GND 4.5V to 24V

PVIN 4.5V to 24V

Electrical Characteristics Specifications with standard typeface are for TJ = 25°C, and those in boldface type

apply over the full Operating Temperature Range (TJ = −40°C to +125°C). Minimum and Maximum limits are guaranteed through

test, design or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C and are provided for

reference purposes only. Unless otherwise specified VIN = 12V.

Symbol Parameter Condition Min Typ Max Units

VFB Feedback Pin Voltage VIN = 4.5V to 24V

ISW = 0A to 3A 1.225 1.254 1.282 V

ICL Switch Current Limit VCBOOT = VSW + 5V 3.6 4.9 6.4 A

RDS_ON Switch On Resistance ISW = 3A 0.13 0.22 Ω

IQOperating Quiescent Current VFB = 1.5V 1.3 2mA

VUVLO AVIN Under Voltage Lockout Rising VIN 3.9 4.125 4.3 V

VUVLO HYS AVIN Under Voltage Lockout Hysteresis 60 120 mV

ISD Shutdown Quiescent Current VSD = 0V 12 25 µA

kON Switch On-Time Constant ION = 50 µA to 100 µA 50 66 82 µA µs

VD ON RON Voltage 0.35 0.65 0.95 V

TOFF_MIN Minimum Off Time FB = 1.24V

FB = 0V 165

250

µs

TON MIN Minimum On-time 400 ns

VEXTVCC EXTVCC Voltage 3.30 3.65 4.00 V

ΔVEXTVCC

EXTVCC Load Regulation IEXTVCC = 0 µA to 50 µA 0.03 0.5 %

VPWRGD PGOOD Threshold (PGOOD Transition

from Low to High)

With respect to VFB 91.5 93.5 95.5 %

VPG_HYS PGOOD Hysteresis 1 2.1 %

IOL PGOOD Low Sink Current VPGOOD = 0.4V 0.5 2 mA

IOH PGOOD High Leakage Current 50 nA

IFB Feedback Pin Bias Current VFB = 1.2V 0 nA

ISS_SOURCE Soft-Start Pin Source Current VSS = 0V 0.7 11.4 µA

ISS SINK Soft-Start Pin Sink Current VSS = 1.2V

VSD = 0V 15 mA

ISD Shutdown Pull-Up Current VSD = 0V 1 3µA

VIH SD Pin Minimum High Input Level 1.8 V

VIL SD Pin Maximum Low Input Level 0.6 V

θJ-A Thermal Resistance 35.1 °C/W

Note 1: Absolute Maximum Ratings indicate limits beyond which damage may occur to the device. Operating Ratings indicate conditions for which the device is

intended to be functional, but do not guarantee specific performance limits. For guaranteed specifications, see Electrical Characteristics.

Note 2: Without PCB copper enhancements. The maximum power dissipation must be derated at elevated temperatures and is limited by TJMAX (maximum

junction temperature), θJ-A (junction to ambient thermal resistance) and TA (ambient temperature). The maximum power dissipation at any temperature is:

PDissMAX = (TJMAX - TA) /θJ-A up to the value listed in the Absolute Maximum Ratings. θJ-A for TSSOP-16 package is 38.1°C/W, TJMAX = 125°C.

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LM2696

Typical Performance Characteristics

IQ vs Temp

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IQ vs VIN

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IQ in Shutdown vs Temp

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IQ vs VIN in Shutdown

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Shutdown Thresholds vs Temp

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EXTVCC vs Temp

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LM2696

EXTVCC vs VIN

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EXTVCC vs Load Current

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Feedback Threshold Voltage vs Temp

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kON vs Temp

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Switch ON Time vs RON Pin Current

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Min Off-Time vs Temp

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LM2696

Max and Min Duty-Cycle vs Freq

(Min TON = 400 ns, Min TOFF = 200 ns)

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FET Resistance vs Temp

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RON Pin Voltage vs Temp

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Current Limit vs Temp

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LM2696

Block Diagram

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Application Information

CONSTANT ON-TIME CONTROL OVERVIEW

The LM2696 buck DC-DC regulator is based on the constant

on-time control scheme. This topology relies on a fixed switch

on-time to regulate the output. The on-time can be set man-

ually by adjusting the size of an external resistor (RON). The

LM2696 automatically adjusts the on-time inversely with the

input voltage (AVIN) to maintain a constant frequency. In con-

tinuous conduction mode (CCM) the frequency depends only

on duty cycle and on-time. This is in contrast to hysteretic

regulators where the switching frequency is determined by

the output inductor and capacitor. In discontinuous conduc-

tion mode (DCM), experienced at light loads, the frequency

will vary according to the load. This leads to high efficiency

and excellent transient response.

The on-time will remain constant for a given VIN unless an

over-current or over-voltage event is encountered. If these

conditions are encountered the FET will turn off for a minimum

pre-determined time period. This minimum TOFF (250 ns) is

internally set and cannot be adjusted. After the TOFF period

has expired the FET will remain off until the comparator trip-

point has been reached. Upon passing this trip-point the FET

will turn back on, and the process will repeat.

Switchers that regulate using an internal comparator to sense

the output voltage for switching decisions, such as hysteretic

or constant on-time, require a minimum ESR. A minimum

ESR is required so that the control signal will be dominated

by ripple that is in phase with the switchpin. Using a control

signal dominated by voltage ripple that is in phase with the

switchpin eliminates the need for compensation, thus reduc-

ing parts count and simplifying design. Alternatively, an RC

feed forward scheme may be used to eliminate the need for

a minimum ESR.

INTERNAL OPERATION

UNDER-VOLTAGE COMPARATOR

An internal comparator is used to monitor the feedback pin for

sensing under-voltage output events. If the output voltage

drops below the UVP threshold the power-good flag will fall.

ON-TIME GENERATOR SHUTDOWN

The on-time for the LM2696 is inversely proportional to the

input voltage. This scheme of on-time control maintains a

constant frequency over the input voltage range. The on-time

can be adjusted by using an external resistor connected be-

tween the PVIN and RON pins.

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LM2696

CURRENT LIMIT

The LM2696 contains an intelligent current limit off-timer. If

the peak current in the internal FET exceeds 4.9A the present

on-time is terminated; this is a cycle-by-cycle current limit.

Following the termination of the on-time, a non-resetable ex-

tended off timer is initiated. The length of the off-time is

proportional to the feedback voltage. When FB = 0V the off-

time is preset to 20 µs. This condition is often a result of in

short circuit operation when a maximum amount of off-time is

required. This amount of time ensures safe short circuit op-

eration up to the maximum input voltage of 24V.

In cases of overload (not complete short circuit, FB > 0V) the

current limit off-time is reduced. Reduction of the off-time dur-

ing smaller overloads reduces the amount of fold back. This

also reduces the initial startup time.

N-CHANNEL HIGH SIDE SWITCH AND DRIVER

The LM2696 utilizes an integrated N-Channel high side

switch and associated floating high voltage gate driver. This

gate driver circuit works in conjunction with an external boot-

strap capacitor and an internal diode. The minimum off-time

(165 ns) is set to ensure that the bootstrap capacitor has suf-

ficient time to charge.

THERMAL SHUTDOWN

An internal thermal sensor is incorporated to monitor the die

temperature. If the die temp exceeds 165ºC then the sensor

will trip causing the part to stop switching. Soft-start will restart

after the temperature falls below 155ºC.

COMPONENT SELECTION

As with any DC-DC converter, numerous trade-offs are

present that allow the designer to optimize a design for effi-

ciency, size and performance. These trade-offs are taken into

consideration throughout this section.

The first calculation for any buck converter is duty cycle. Ig-

noring voltage drops associated with parasitic resistances

and non-ideal components, the duty cycle may be expressed

as:

A duty cycle relationship that considers the voltage drop

across the internal FET and voltage drop across the external

catch diode may be expressed as:

Where VD is the forward voltage of the external catch diode

(DCATCH) and VSW is the voltage drop across the internal FET.

FREQUENCY SELECTION

Switching frequency affects the selection of the output induc-

tor, capacitor, and overall efficiency. The trade-offs in fre-

quency selection may be summarized as; higher switching

frequencies permit use of smaller inductors possibly saving

board space at the trade-off of lower efficiency. It is recom-

mended that a nominal frequency of 300 kHz should be used

in the initial stages of design and iterated if necessary.

The switching frequency of the LM2696 is set by the resistor

connected to the RON pin. This resistor controls the current

flowing into the RON pin and is directly related to the on-time

pulse. Connecting a resistor from this pin to PVIN allows the

switching frequency to remain constant as the input voltage

changes. In normal operation this pin is approximately 0.65V

above GND. In shutdown, this pin becomes a high impedance

node to prevent current flow.

The ON time may be expressed as:

Where VIN is the voltage at the high side of the RON resistor

(typically PVIN), VD is the diode voltage present at the RON

pin (0.65V typical), RON is in kΩ, and kON is a constant value

set internally (66 µA•µs nominal). This equation can be re-

arranged such that RON is a function of switching frequency:

Where fSW is in kHz.

In CCM the frequency may be determined using the relation-

ship:

(TON is in µs)

Which is typically used to set the switching frequency.

Under no condition should a bypass capacitor be connected

to the RON pin. Doing so couples any AC perturbations into

the pin and prevents proper operation.

INDUCTOR SELECTION

Selecting an inductor is a process that may require several

iterations. The reason for this is that the size of the inductor

influences the amount of ripple present at the output that is

critical to the stability of an adaptive on-time circuit. Typically,

an inductor is selected such that the maximum peak-to-peak

ripple current is equal to 30% of the maximum load current.

The inductor current ripple (ΔIL) may be expressed as:

Therefore, L can be initially set to the following by applying

the 30% guideline:

The other features of the inductor that should be taken into

account are saturation current and core material. A shielded

inductor or low profile unshielded inductor is recommended

to reduce EMI.

OUTPUT CAPACITOR

The output capacitor size and ESR have a direct affect on the

stability of the loop. This is because the adaptive on-time

control scheme works by sensing the output voltage ripple

and switching appropriately. The output voltage ripple on a

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LM2696

buck converter can be approximated by assuming that the AC

inductor ripple current flows entirely into the output capacitor

and the ESR of the capacitor creates the voltage ripple. This

is expressed as:

ΔVOUT≈ ΔIL • RESR

To ensure stability, two constraints need to be met. These

constraints are the voltage ripple at the feedback pin must be

greater than some minimum value and the voltage ripple must

be in phase with the switch pin.

The ripple voltage necessary at the feedback pin may be es-

timated using the following relationship:

ΔVFB > −0.057 • fSW + 35

Where fSW is in kHz and ΔVFB is in mV.

This minimum ripple voltage is necessary in order for the

comparator to initiate switching. The voltage ripple at the

feedback pin must be in-phase with the switch. Because the

ripple due to the capacitor charging and capacitor ESR are

out of phase, the ripple due to capacitor ESR must dominate.

The ripple at the output may be calculated by multiplying the

feedback ripple voltage by the gain seen through the feed-

back resistors. This gain H may be expressed as:

To simplify design and eliminate the need for high ESR output

capacitors, an RC network may be used to feed forward a

signal from the switchpin to the feedback (FB) pin. See the

‘Ripple Feed Forward’ section for more details.

Typically, the best performance is obtained using POSCAPs,

SP CAPs, tantalum, Niobium Oxide, or similar chemistry type

capacitors. Low ESR ceramic capacitors may be used in con-

junction with the RC feed forward scheme; however, the feed

forward voltage at the feedback pin must be greater than 30

mV.

RIPPLE FEED FORWARD

An RC network may be used to eliminate the need for high

ESR capacitors. Such a network is connected as shown in

Figure 1.

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FIGURE 1. RC Feed Forward Network

The value of Rff should be large in order to prevent any po-

tential offset in VOUT. Typically the value of Rff is on the order

of 1 MΩ and the value of RFB1 should be less than 10 kΩ. The

large difference in resistor values minimizes output voltage

offset errors in DCM. The value of the capacitor may be se-

lected using the following relationship:

Where the on-time (TON_MIN) is in µs, and the resistance (Rff)

is in MΩ.

FEEDBACK RESISTORS

The feedback resistors are used to scale the output voltage

to the internal reference value such that the loop can be reg-

ulated. The feedback resistors should not be made arbitrarily

large as this creates a high impedance node at the feedback

pin that is more susceptible to noise. Typically, RFB2 is on the

order of 1 kΩ. To calculate the value of RFB1, one may use

the relationship:

Where VFB is the internal reference voltage that can be found

in the electrical characteristics table (1.254V typical).

The output voltage value can be set in a precise manner by

taking into account the fact that the reference voltage is reg-

ulating the bottom of the output ripple as opposed to the

average value. This relationship is shown in Figure 2.

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LM2696

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FIGURE 2. Average and Ripple Output Voltages

It can be seen that the average output voltage is higher than

the gained up reference by exactly half the output voltage rip-

ple. The output voltage may then be appended according to

the voltage ripple. The appended VOUT term may be ex-

pressed using the relationship:

One should note that for high output voltages (>5V), a load of

approximately 15 mA may be required for the output voltage

to reach the desired value.

INPUT CAPACITOR

Because PVIN is the power rail from which the output voltage

is derived, the input capacitor is typically selected according

to the load current. In general, package size and ESR deter-

mine the current capacity of a capacitor. If these criteria are

met, there should be enough capacitance to prevent

impedance interactions with the source. In general, it is rec-

ommended to use a low ESR, high capacitance electrolytic

and ceramic capacitor in parallel. Using two capacitors in par-

allel ensures adequate capacitance and low ESR over the

operating range. The Sanyo MV-WX series electrolytic ca-

pacitors and a ceramic capacitor with X5R or X7R dielectric

are an excellent combination. To calculate the input capacitor

RMS, one may use the following relationship:

that can be approximated by,

Typical values are 470 µF for the electrolytic capacitor and

0.1 µF for the ceramic capacitor.

AVIN CAPACITOR

AVIN is the analog bias rail of the device. It should be by-

passed externally with a small (1 µF) ceramic capacitor to

prevent unwanted noise from entering the device. In a shut-

down state the current needed by AVIN will drop to approxi-

mately 12 µA, providing a low power sleep state.

In most cases of operation, AVIN is connected to PVIN; how-

ever, it is possible to have split rail operation where AVIN is at

a higher voltage than PVIN. AVIN should never be lower than

PVIN. Splitting the rails allows the power conversion to occur

from a lower rail than the AVIN operating range.

SOFT-START CAPACITOR

The SS capacitor is used to slowly ramp the reference from

0V to its final value of 1.25V (during shutdown this pin will be

discharged to 0V). This controlled startup ability eliminates

large in-rush currents in an attempt to charge up the output

capacitor. By changing the value of this capacitor, the dura-

tion of the startup may be changed accordingly. The startup

time may be calculated using the following relationship:

Where ISS is the soft-start pin source current (1 µA typical)

that may be found in the electrical characteristics table. While

the CSS capacitor can be sized to meet the startup require-

ments, there are limitations to its size. If the capacitor is too

small, the soft-start will have little effect as the reference volt-

age is rising faster than the output capacitor can be charged

causing the part to go into current limit. Therefore a minimum

soft-start time should be taken into account. This can be de-

termined by:

While COUT and VOUT control the slew rate of the output volt-

age, the total amount of time the LM2696 takes to startup is

dependent on two other terms. See the “Startup” section for

more information.

EXTVCC CAPACITOR

External VCC is a 3.65V rail generated by an internal sub-reg-

ulator that powers the parts internal circuitry. This rail should

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LM2696

be bypassed with a 1 µF ceramic capacitor (X5R or equivalent

dielectric). Although EXTVCC is for internal use, it can be used

as an external rail for extremely light loads (<50 µA). If

EXTVCC is accidentally shorted to GND the part is protected

by a 5 mA current limit. This rail also has an under-voltage

lockout that will prevent the part from switching if the

EXTVCC voltage drops.

SHUTDOWN

The state of the shutdown pin enables the device or places it

in a sleep state. This pin has an internal pull-up and may be

left floating or connected to a high logic level. Connecting this

pin to GND will shutdown the part. Shutting down the part will

prevent the part from switching and reduce the quiescent cur-

rent drawn by the part. This pin must be bypassed with a 1 nF

ceramic capacitor (X5R or Y5V) to ensure proper logic thresh-

olds.

CBOOT CAPACITOR

The purpose of an external bootstrap capacitor is to turn the

FET on by using the SW node as a pedestal. This allows the

voltage on the CBOOT pin to be greater than VIN. Whenever

the catch diode is conducting and the SW node is at GND, an

internal diode will conduct that charges the CBOOT capacitor

to approximately 4V. When the SW node rises, the CBOOT

pin will rise to approximately 4V above the SW node. For op-

timal performance, a 0.1 µF ceramic capacitor (X5R or equiv-

alent dielectric) should be used.

PGOOD RESISTOR

The PGOOD resistor is used to pull the PGOOD pin high

whenever a steady state operating range is achieved. This

resistor needs to be sized to prevent excessive current from

flowing into the PGOOD pin whenever the open drain FET is

turned on. The recommendation is to use a 10 kΩ–100 kΩ

resistor. This range of values is a compromise between rise

time and power dissipation.

CATCH DIODE

The catch or freewheeling diode acts as the bottom switch in

a non-synchronous buck switcher. Because of this, the diode

has to handle the full output current whenever the FET is not

conducting. Therefore, it must be sized appropriately to han-

dle the current. The average current through the diode can be

calculated by the equation:

ID_AVG = IOUT•(1–D)

Care should also be taken to ensure that the reverse voltage

rating of the diode is adequate. Whenever the FET is con-

ducting the voltage across the diode will be approximately

equal to VIN. It is recommended to have a reverse rating that

is equal to 120% of VIN to ensure adequate guard banding

against any ringing that could occur on the switch node.

Selection of the catch diode is critical to overall switcher per-

formance. To obtain the optimal performance, a Schottky

diode should be used due to their low forward voltage drop

and fast recovery.

BYPASS CAPACITOR

A bypass capacitor must be used on the AVIN line to help de-

couple any noise that could interfere with the analog circuitry.

Typically, a small (1 µF) ceramic capacitor is placed as close

as possible to the AVIN pin.

EXTERNAL OPERATION STARTUP

The total startup time, from the initial VIN rise to the time

VOUT reaches its nominal value is determined by three sepa-

rate steps. Upon the rise of VIN, the first step to occur is that

the EXTVCC voltage has to reach its nominal output voltage

of 3.65V before the internal circuitry is active. This time is dic-

tated by the output capacitance (1 µF) and the current limit of

the regulator (5 mA typical), which will always be on the order

of 730 µs. Upon reaching its steady state value, an internal

delay of 200 µs will occur to ensure stable operation. Upon

completion the LM2696 will begin switching and the output

will rise. The rise time of the output will be governed by the

soft-start capacitor. To highlight these three steps a timing

diagram please refer to Figure 3.

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LM2696

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FIGURE 3. Startup Timing Diagram

UNDER- & OVER-VOLTAGE CONDITIONS

The LM2696 has a built in under-voltage comparator that

controls PGOOD. Whenever the output voltage drops below

the set threshold, the PGOOD open drain FET will turn on

pulling the pin to ground. For an over-voltage event, there is

no separate comparator to control PGOOD. However, the

loop responds to prevent this event from occurring because

the error comparator is essentially sensing an OVP event. If

the output is above the feedback threshold then the part will

not switch back on; therefore, the worst-case condition is one

on-time pulse.

CURRENT LIMIT

The LM2696 utilizes a peak-detect current limit that senses

the current through the FET when conducting and will imme-

diately terminate the on-pulse whenever the peak current

exceeds the threshold (4.9A typical). In addition to terminating

the present on-pulse, it enforces a mandatory off-time that is

related to the feedback voltage.

If current limit trips and the feedback voltage is close to its

nominal value of 1.25V, the off-time imposed will be relatively

short. This is to prevent the output from dropping or any fold

back from occurring if a momentary short occurred because

of a transient or load glitch. If a short circuit were present, the

off-time would extend to approximately 12 µs. This ensures

that the inductor current will reach a low value (approximately

0A) before the next switching cycle occurs. The extended off-

time prevents runaway conditions caused by hard shorts and

high side blanking times.

If the part is in an over current condition, the output voltage

will begin to drop as shown in Figure 4. If the output voltage

is dropping and the current is below the current limit threshold,

(I1), the part will assert a pulse (t2) after a minimum off-time

(t1). This is in an attempt to raise the output voltage.

If the part is in an over current condition and the output voltage

is below the regulation value (VL) as shown in Figure 4, the

part will assert a pulse of minimal width (t4) and extend the

off-time (t5). In the event that the voltage is below the regula-

tion value (VL) and the current is below the current limit value,

the part will assert two (or more) pulses separated by some

minimal off-time (t1).

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LM2696

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FIGURE 4. Fault Condition Timing

Legend:

t1: Min off-time (165 ns typical)

t2: On-time (set by the user)

t3: Min off-time (165 ns typical)

t4: Blanking time (165 ns typical)

t5: Extended off-time (12 µs typical)

VL: UVP threshold

The last benefit of this scheme is when the short circuit is

removed, and full load is re-applied, the part will automatically

recover into the load. The variation in the off-time removes

the constraints of other frequency fold back systems where

the load would typically have to be reduced.

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FIGURE 5. Normalized Output Voltage

Versus Load Current

MODES OF OPERATION

Since the LM2696 utilizes a catch diode, whenever the load

current is reduced to a point where the inductor ripple is

greater than two times the load current, the part will enter dis-

continuous operation. This is because the diode does not

permit the inductor current to reverse direction. The point at

which this occurs is the critical conduction boundary and can

be calculated by the following equation:

One advantage of the adaptive on-time control scheme is that

during discontinuous conduction mode the frequency will

gradually decrease as the load current decreases. In DCM

the switching frequency may be determined using the rela-

tionship:

It can be seen that there will always be some minimum switch-

ing frequency. The minimum switching frequency is deter-

mined by the parameters above and the minimum load

presented by the feedback resistors. If there is some mini-

mum frequency of operation the feedback resistors may be

sized accordingly.

The adaptive on-time control scheme is effectively a pulse-

skipping mode, but since it is not tied directly to an internal

clock, its pulse will only occur when needed. This is in contrast

to schemes that synchronize to a reference clock frequency.

The constant on-time pulse-skipping/DCM mode minimizes

output voltage ripple and maximizes efficiency.

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LM2696

Several diagrams are shown in Figure 6 illustrating continu-

ous conduction mode (CCM), discontinuous conduction

mode (DCM), and the boundary condition.

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FIGURE 6. Modes of Operation

It can be seen that in DCM, whenever the inductor runs dry

the SW node will become high impedance. Ringing will occur

as a result of the LC tank circuit formed by the inductor and

the parasitic capacitance at the SW node.

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FIGURE 7. Parasitic Tank Circuit at the Switchpin

LINE REGULATION

The LM2696 regulates to the lowest point of the output volt-

age (VL in Figure 8 ). This is to say that the output voltage may

be represented by a waveform that is some average voltage

with ripple. The LM2696 will regulate to the trough of the rip-

ple.

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LM2696

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FIGURE 8. Average Output Voltage and Regulation Point

The output voltage is given by the following relationship:

as discussed in the Feedback Resistor section of this docu-

ment.

TRANSIENT RESPONSE

Constant on-time architectures have inherently excellent tran-

sient line and load response. This is because the control loop

is extremely fast. Any change in the line or load conditions will

result in a nearly instantaneous response in the PWM off time.

If one considers the switcher response to be nearly cycle-by-

cycle, and amount of energy contained in a single PWM pulse,

there will be very little change in the output for a given change

in the line or load.

EFFICIENCY

The constant on-time architecture features high efficiency

even at light loads. The ability to achieve high efficiency at

light loads is due to the fact that the off-time will become nec-

essarily long at light loads. Having extended the off-time,

there is little mechanism for loss during this interval.

The efficiency is easily estimated using the following relation-

ships:

Power loss due to FET:

PFET = PC + PGC + PSW

Where:

PC = D • (IOUT2 • RDS_ON)

PGC = AVIN + VGS • QGS • fSW

PSW = 0.5 • VIN · IOUT • (tr + tf) • fSW

Typical values are:

RDS_ON = 130 mΩ

VGS = 4V

QGS = 13.3 nC

tr = 3.8 ns

tf = 4.5 ns

Power loss due to catch diode:

PD = (1-D) • (IOUT • Vf)

Power loss due to DCR and ESR:

PDCR = IOUT2 • RDCR

PESR_OUTPUT = IRIPPLE2/√12 • RESR_OUTPUT

PESR_INPUT = IOUT2(D(1-D)) • RESR_INPUT

Power loss due to Controller:

PCONT = VIN • IQ

IQ is typically 1.3 mA

The efficiency may be calculated as shown below:

Total power loss = PFET + PD + PDCR + PESR_OUTPUT + PESR_IN-

PUT + PCONT

Power Out = IOUT • VOUT

PRE-BIAS LOAD STARTUP

Should the LM2696 start into a pre-biased load the output will

not be pulled low. This is because the part is asynchronous

and cannot sink current. The part will respond to a pre-biased

load by simply enabling PWM high or extending the off-time

until regulation is achieved. This is to say that if the output

voltage is greater than the regulation voltage the off-time will

extend until the voltage discharges through the feedback re-

sistors. If the load voltage is greater than the regulation volt-

age, a series of pulses will charge the output capacitor to its

regulation voltage.

THERMAL CONSIDERATIONS

The thermal characteristics of the LM2696 are specified using

the parameter θJA, which relates the junction temperature to

the ambient temperature. While the value of θJA is specific to

a given set of test parameters (including board thickness,

number of layers, orientation, etc), it provides the user with a

common point of reference.

To obtain an estimate of a devices junction temperature, one

may use the following relationship:

TJ = PIN (1-Efficiency) x θJA + TA

Where:

TJ is the junction temperature in ºC

PIN is the input power in Watts (PIN = VIN·IIN)

θJA is the thermal coefficient of the LM2696

TA is the ambient temperature in ºC

LAYOUT CONSIDERATIONS

The LM2696 regulation and under-voltage comparators are

very fast and will respond to short duration noise pulses. Lay-

out considerations are therefore critical for optimum perfor-

15 www.national.com

LM2696

mance. The components at pins 5, 6, 7, 12 and 13 should be

as physically close as possible to the IC, thereby minimizing

noise pickup in the PC traces. If the internal dissipation of the

LM2696 produces excessive junction temperatures during

normal operation, good use of the PC board’s ground plane

can help considerably to dissipate heat. The exposed pad on

the bottom of the TSSOP-16 package can be soldered to a

ground plane on the PC board, and that plane should extend

out from beneath the IC to help dissipate the heat. Use of

several vias beneath the part is also an effective method of

conducting heat. Additionally, the use of wide PC board

traces, where possible, can also help conduct heat away from

the IC. Judicious positioning of the PC board within the end

product, along with use of any available air flow (forced or

natural convection) can help reduce the junction tempera-

tures. Traces in the power plane (Figure 9) should be short

and wide to minimize the trace impedance; they should also

occupy the smallest area that is reasonable to minimize EMI.

Sizing the power plane traces is a tradeoff between current

capacity, inductance, and thermal dissipation. For more in-

formation on layout considerations, please refer to National

Semiconductor Application Note AN-1229.

20153450

FIGURE 9. Bold Traces Are In The Power Plane

www.national.com 16

LM2696

20153451

FIGURE 10. 5V-to-2.5V Voltage Applications Circuit

Bill of Materials

(Figure 10: Medium Voltage Board, 5V-to-2.5V conversion, fsw = 300 kHz)

Designator Function Description Vendor Part Number

CIN Input Cap 470 µF Sanyo 10MV470WX

CBY Bypass Cap 0.1 µF Vishay VJ0805Y104KXAM

CSS Soft-Start Cap 0.01 µF Vishay VJ080JY103KXX

CEXT EXTVCC 1 µF Vishay VJ0805Y105JXACW1BC

CBOOT Boot 0.1 µF Vishay VJ0805Y104KXAM

CAVIN Analog VIN 1 µF Vishay VJ0805Y105JXACW1BC

COUT Output Cap 47 µF AVX TPSW476M010R0150

CSD Shutdown Cap 1 nF Vishay VJ0805Y102KXXA

RFB1 High Side FB Res 1 kΩVishay CRCW08051001F

RFB2 Low Side RB Res 1 kΩVishay CRCW08051001F

RON On Time Res 143 kΩVishay CRCW08051433F

DCATCH Boot Diode 40V @ 3A Diode Central Semi CMSH3-40M-NST

L Output Inductor 6.8 uH, 4.9A ISAT Coilcraft MSS1260-682MX

17 www.national.com

LM2696

20153452

FIGURE 11. 12V-to 3.3V Voltage Applications Circuit

Bill of Materials

(Figure 11: Medium Voltage Board, 12V-to-3.3V conversion, fsw = 300 kHz)

Designator Function Description Vendor Part Number

CIN Input Cap 560 µF Sanyo 35MV560WX

CBY Bypass Cap 0.1 µF Vishay VJ0805Y104KXAM

CSS Soft-Start Cap 0.01 µF Vishay VJ080JY103KXX

CEXT EXTVCC 1 µF Vishay VJ0805Y105JXACW1BC

CBOOT Boot 0.1 µF Vishay VJ0805Y104KXAM

CAVIN Analog VIN 1 µF Vishay VJ0805Y105JXACW1BC

COUT Output Cap 100 µF Sanyo 6SVPC100M

CSD Shutdown Cap 1 nF Vishay VJ0805Y102KXXA

Cff Feedforward Cap 560 pF Vishay VJ0805A561KXXA

Rff Feedforward Res 1 MΩVishay CRCW08051004F

RFB1 High Side FB Res 1.62 kΩVishay CRCW08051621F

RFB2 Low Side RB Res 1 kΩVishay CRCW08051001F

RON On Time Res 143 kΩVishay CRCW08051433F

DCATCH Boot Diode 40V @ 3A Diode Central Semi CMSH3-40M-NST

L Output Inductor 10 uH, 5.4A ISAT Coilcraft MSS1278-103MX

www.national.com 18

LM2696

Physical Dimensions inches (millimeters) unless otherwise noted

eTSSOP-16 Package

Order Number LM2696MXA or LM2696MXAX

NS Package Number MXA16A

19 www.national.com

LM2696

Notes

LM2696 3A, Constant On Time Buck Regulator

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