LM348MXNOPB - NATIONAL SEMICONDUCTOR

LM3488,LM3488Q

LM3488/LM3488Q High Efficiency Low-Side N-Channel Controller for Switching

Regulators

Literature Number: SNVS089L

LM3488/LM3488Q

October 17, 2011

High Efficiency Low-Side N-Channel Controller for

Switching Regulators

General Description

The LM3488 is a versatile Low-Side N-FET high performance

controller for switching regulators. It is suitable for use in

topologies requiring low side FET, such as boost, flyback,

SEPIC, etc. Moreover, the LM3488 can be operated at ex-

tremely high switching frequency in order to reduce the overall

solution size. The switching frequency of LM3488 can be ad-

justed to any value between 100kHz and 1MHz by using a

single external resistor or by synchronizing it to an external

clock. Current mode control provides superior bandwidth and

transient response, besides cycle-by-cycle current limiting.

Output current can be programmed with a single external re-

sistor.

The LM3488 has built in features such as thermal shutdown,

short-circuit protection and over voltage protection. Power

saving shutdown mode reduces the total supply current to

5µA and allows power supply sequencing. Internal soft-start

limits the inrush current at start-up.

Key Specifications

■Wide supply voltage range of 2.97V to 40V

■100kHz to 1MHz Adjustable and Synchronizable clock

frequency

■±1.5% (over temperature) internal reference

■5µA shutdown current (over temperature)

Features

■LM3488Q is AEC-Q100 qualified and manufactured on an

Automotive Grade Flow

■8-lead Mini-SO8 (MSOP-8) package

■Internal push-pull driver with 1A peak current capability

■Current limit and thermal shutdown

■Frequency compensation optimized with a capacitor and

a resistor

■Internal softstart

■Current Mode Operation

■Undervoltage Lockout with hysteresis

Applications

■Distributed Power Systems

■Notebook, PDA, Digital Camera, and other Portable

Applications

■Offline Power Supplies

■Set-Top Boxes

Typical Application Circuit

10138844

Typical SEPIC Converter

LM3488/LM3488Q High Efficiency Low-Side N-Channel Controller for Switching Regulators

Connection Diagram

10138802

8 Lead Mini SO8 Package (MSOP-8 Package)

Package Marking and Ordering Information

Order Number Package Type Package Marking Supplied As Feature

LM3488MM

MSOP-8

S21B 1000 units on Tape and Reel

LM3488MMX 3500 units on Tape and Reel

LM3488QMM

SSKB

1000 units on Tape and Reel AEC-Q100 (Grade 1) qualified.

Automotive Grade Production

Flow*

LM3488QMMX 3500 units on Tape and Reel

*Automotive Grade (Q) product incorporates enhanced manufacturing and support processes for the automotive market, including defect detection methodologies.

Reliability qualification is compliant with the requirements and temperature grades defined in the AEC-Q100 standard. Automotive grade products are identified

with the letter Q. For more information go to http://www.national.com/automotive.

Pin Descriptions

Pin Name Pin Number Description

ISEN 1Current sense input pin. Voltage generated across an external sense

resistor is fed into this pin.

COMP 2 Compensation pin. A resistor, capacitor combination connected to this

pin provides compensation for the control loop.

FB 3 Feedback pin. The output voltage should be adjusted using a resistor

divider to provide 1.26V at this pin.

AGND 4 Analog ground pin.

PGND 5 Power ground pin.

DR 6 Drive pin of the IC. The gate of the external MOSFET should be

connected to this pin.

FA/SYNC/SD 7 Frequency adjust, synchronization, and Shutdown pin. A resistor

connected to this pin sets the oscillator frequency. An external clock

signal at this pin will synchronize the controller to the frequency of the

clock. A high level on this pin for ≥ 30µs will turn the device off. The

device will then draw less than 10µA from the supply.

VIN 8 Power supply input pin.

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LM3488/LM3488Q

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

Input Voltage 45V

FB Pin Voltage -0.4V < VFB < 7V

FA/SYNC/SD Pin Voltage -0.4V < VFA/SYNC/SD

< 7V

Peak Driver Output Current (<10µs) 1.0A

Power Dissipation Internally Limited

Storage Temperature Range −65°C to +150°C

Junction Temperature +150°C

ESD Susceptibilty

Human Body Model (Note 2) 2kV

Lead Temperature

MM Package

Vapor Phase (60 sec.)

Infared (15 sec.)

215°C

260°C

DR Pin Voltage −0.4V ≤ VDR ≤ 8V

ILIM Pin Voltage 600mV

Operating Ratings (Note 1)

Supply Voltage 2.97V ≤ VIN ≤ 40V

Junction Temperature

Range −40°C ≤ TJ ≤ +125°C

Switching Frequency 100kHz ≤ FSW ≤ 1MHz

Electrical Characteristics

Specifications in Standard type face are for TJ = 25°C, and in bold type face apply over the full Operating Temperature

Range. Unless otherwise specified, VIN = 12V, RFA = 40kΩ

Symbol Parameter Conditions Typical Limit Units

VFB Feedback Voltage VCOMP = 1.4V,

2.97 ≤ VIN ≤ 40V

1.26

1.2507/1.24

1.2753/1.28

V(min)

V(max)

ΔVLINE Feedback Voltage Line

Regulation

2.97 ≤ VIN ≤ 40V 0.001

%/V

ΔVLOAD Output Voltage Load

Regulation

IEAO Source/Sink ±0.5

%/V (max)

VUVLO Input Undervoltage Lock-out 2.85

2.97

V(max)

VUV(HYS) Input Undervoltage Lock-out

Hysteresis

170

130

210

mV (min)

mV (max)

Fnom Nominal Switching Frequency RFA = 40KΩ400

360

430

kHz

kHz(min)

kHz(max)

RDS1 (ON) Driver Switch On Resistance

(top)

IDR = 0.2A, VIN= 5V 16 Ω

RDS2 (ON) Driver Switch On Resistance

(bottom)

IDR = 0.2A 4.5 Ω

VDR (max) Maximum Drive Voltage

Swing(Note 6)

VIN < 7.2V VIN V

VIN ≥ 7.2V 7.2

Dmax Maximum Duty Cycle(Note 7) 100 %

Tmin (on) Minimum On Time 325

230

550

nsec

nsec(min)

nsec(max)

ISUPPLY Supply Current (switching) (Note 9)2.7 3.0

mA (max)

IQQuiescent Current in

Shutdown Mode

VFA/SYNC/SD = 5V(Note 10), VIN =

µA

µA (max)

VSENSE Current Sense Threshold

Voltage

VIN = 5V 156

135/ 125

180/ 190

mV (min)

mV (max)

VSC Short-Circuit Current Limit

Sense Voltage

VIN = 5V 343

250

415

mV (min)

mV (max)

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LM3488/LM3488Q

Symbol Parameter Conditions Typical Limit Units

VSL Internal Compensation Ramp

Voltage

VIN = 5V 92

132

mV(min)

mV(max)

VSL ratio VSL/VSENSE 0.49 0.30

0.70

(min)

(max)

VOVP Output Over-voltage

Protection (with respect to

feedback voltage) (Note 8)

VCOMP = 1.4V 50

32/ 25

78/ 85

mV(min)

mV(max)

VOVP(HYS) Output Over-Voltage

Protection Hysteresis(Note 8)

VCOMP = 1.4V 60

110

mV(min)

mV(max)

Gm Error Ampifier

Transconductance

VCOMP = 1.4V

IEAO = 100µA (Source/Sink)

800

600/ 365

1000/ 1265

µmho

µmho (min)

µmho (max)

AVOL Error Amplifier Voltage Gain VCOMP = 1.4V

IEAO = 100µA (Source/Sink)

V/V

V/V (min)

V/V (max)

IEAO Error Amplifier Output Current

(Source/ Sink)

Source, VCOMP = 1.4V, VFB = 0V 110

80/ 50

140/ 180

µA

µA (min)

µA (max)

Sink, VCOMP = 1.4V, VFB = 1.4V −140

−100/ −85

−180/ −185

µA

µA (min)

µA (max)

VEAO Error Amplifier Output Voltage

Swing

Upper Limit

VFB = 0V

COMP Pin = Floating

2.2

1.8

2.4

V(min)

V(max)

Lower Limit

VFB = 1.4V

0.56

0.2

1.0

V(min)

V(max)

TSS Internal Soft-Start Delay VFB = 1.2V, VCOMP = Floating 4 msec

TrDrive Pin Rise Time Cgs = 3000pf, VDR = 0 to 3V 25 ns

TfDrive Pin Fall Time Cgs = 3000pf, VDR = 0 to 3V 25 ns

VSD Shutdown and

Synchronization signal

threshold (Note 5)

Output = High 1.27

1.4

V (max)

Output = Low 0.65

0.3

V (min)

ISD Shutdown Pin Current VSD = 5V −1 µA

VSD = 0V +1

IFB Feedback Pin Current 15 nA

TSD Thermal Shutdown 165 °C

Tsh Thermal Shutdown Hysteresis 10 °C

θJA Thermal Resistance MM Package 200 °C/W

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LM3488/LM3488Q

Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the

device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics.

Note 2: The human body model is a 100 pF capacitor discharged through a 1.5kΩ resistor into each pin.

Note 3: All limits are guaranteed at room temperature (standard type face) and at temperature extremes (bold type face). All room temperature limits are 100%

tested. All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods. All limits are used to calculate

Average Outgoing Quality Level (AOQL).

Note 4: Typical numbers are at 25°C and represent the most likely norm.

Note 5: The FA/SYNC/SD pin should be pulled to VIN through a resistor to turn the regulator off.

Note 6: The voltage on the drive pin, VDR is equal to the input voltage when input voltage is less than 7.2V. VDR is equal to 7.2V when the input voltage is greater

than or equal to 7.2V.

Note 7: The limits for the maximum duty cycle can not be specified since the part does not permit less than 100% maximum duty cycle operation.

Note 8: The over-voltage protection is specified with respect to the feedback voltage. This is because the over-voltage protection tracks the feedback voltage.

The over-voltage thresold can be calculated by adding the feedback voltage, VFB to the over-voltage protection specification.

Note 9: For this test, the FA/SYNC/SD Pin is pulled to ground using a 40K resistor .

Note 10: For this test, the FA/SYNC/SD Pin is pulled to 5V using a 40K resistor.

Typical Performance Characteristics Unless otherwise specified, VIN = 12V, TJ = 25°C.

IQ vs Temperature & Input Voltage

10138803

ISupply vs Input Voltage (Non-Switching)

10138834

ISupply vs VIN

10138835

Switching Frequency vs RFA

10138804

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LM3488/LM3488Q

Frequency vs Temperature

10138854

Drive Voltage vs Input Voltage

10138805

Current Sense Threshold vs Input Voltage

10138845

COMP Pin Voltage vs Load Current

10138862

Efficiency vs Load Current (3.3V In and 12V Out)

10138859

Efficiency vs Load Current (5V In and 12V Out)

10138858

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LM3488/LM3488Q

Efficiency vs Load Current (9V In and 12V Out)

10138860

Efficiency vs Load Current (3.3V In and 5V Out)

10138853

Error Amplifier Gain

10138855

Error Amplifier Phase

10138856

COMP Pin Source Current vs Temperature

10138836

Short Circuit Protection vs Input Voltage

10138857

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LM3488/LM3488Q

Compensation Ramp vs Compensation Resistor

10138851

Shutdown Threshold Hysteresis vs Temperature

10138846

Current Sense Voltage vs Duty Cycle

10138852

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LM3488/LM3488Q

Functional Block Diagram

10138806

Functional Description

The LM3488 uses a fixed frequency, Pulse Width Modulated

(PWM), current mode control architecture. In a typical appli-

cation circuit, the peak current through the external MOSFET

is sensed through an external sense resistor. The voltage

across this resistor is fed into the ISEN pin. This voltage is then

level shifted and fed into the positive input of the PWM com-

parator. The output voltage is also sensed through an external

feedback resistor divider network and fed into the error am-

plifier negative input (feedback pin, FB). The output of the

error amplifier (COMP pin) is added to the slope compensa-

tion ramp and fed into the negative input of the PWM com-

parator.

At the start of any switching cycle, the oscillator sets the RS

latch using the SET/Blank-out and switch logic blocks. This

forces a high signal on the DR pin (gate of the external MOS-

FET) and the external MOSFET turns on. When the voltage

on the positive input of the PWM comparator exceeds the

negative input, the RS latch is reset and the external MOSFET

turns off.

The voltage sensed across the sense resistor generally con-

tains spurious noise spikes, as shown in Figure 1. These

spikes can force the PWM comparator to reset the RS latch

prematurely. To prevent these spikes from resetting the latch,

a blank-out circuit inside the IC prevents the PWM comparator

from resetting the latch for a short duration after the latch is

set. This duration is about 150ns and is called the blank-out

time.

Under extremely light load or no-load conditions, the energy

delivered to the output capacitor when the external MOSFET

is on during the blank-out time is more than what is delivered

to the load. An over-voltage comparator inside the LM3488

prevents the output voltage from rising under these condi-

tions. The over-voltage comparator senses the feedback (FB

pin) voltage and resets the RS latch under these conditions.

The latch remains in reset state till the output decays to the

nominal value.

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LM3488/LM3488Q

10138807

FIGURE 1. Basic Operation of the PWM comparator

SLOPE COMPENSATION RAMP

The LM3488 uses a current mode control scheme. The main

advantages of current mode control are inherent cycle-by-cy-

cle current limit for the switch, and simpler control loop char-

acteristics. It is also easy to parallel power stages using

current mode control since as current sharing is automatic.

Current mode control has an inherent instability for duty cy-

cles greater than 50%, as shown in Figure 2. In Figure 2, a

small increase in the load current causes the switch current

to increase by ΔIO. The effect of this load change, ΔI1, is :

From the above equation, when D > 0.5, ΔI1 will be greater

than ΔIO. In other words, the disturbance is divergent. So a

very small perturbation in the load will cause the disturbance

to increase.

To prevent the sub-harmonic oscillations, a compensation

ramp is added to the control signal, as shown in Figure 3.

With the compensation ramp,

10138809

FIGURE 2. Sub-Harmonic Oscillation for D>0.5

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LM3488/LM3488Q

10138811

FIGURE 3. Compensation Ramp Avoids Sub-Harmonic Oscillation

The compensation ramp has been added internally in

LM3488. The slope of this compensation ramp has been se-

lected to satisfy most of the applications. The slope of the

internal compensation ramp depends on the frequency. This

slope can be calculated using the formula:

MC = VSL.FS Volts/second

In the above equation, VSL is the amplitude of the internal

compensation ramp. Limits for VSL have been specified in the

electrical characteristics.

In order to provide the user additional flexibility, a patented

scheme has been implemented inside the IC to increase the

slope of the compensation ramp externally, if the need arises.

Adding a single external resistor, RSL(as shown in Figure 4)

increases the slope of the compensation ramp, MC by :

In this equation, ΔVSL is equal to 40.10-6RSL. Hence,

ΔVSL versus RSL has been plotted in Figure 5 for different fre-

quencies.

10138813

FIGURE 4. Increasing the Slope of the Compensation Ramp

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LM3488/LM3488Q

10138851

FIGURE 5. ΔVSL vs RSL

FREQUENCY ADJUST/SYNCHRONIZATION/SHUTDOWN

The switching frequency of LM3488 can be adjusted between

100kHz and 1MHz using a single external resistor. This re-

sistor must be connected between FA/SYNC/SD pin and

ground, as shown in Figure 6. Please refer to the typical per-

formance characteristics to determine the value of the resistor

required for a desired switching frequency.

The LM3488 can be synchronized to an external clock. The

external clock must be connected to the FA/SYNC/SD pin

through a resistor, RSYNC as shown in Figure 7. The value of

this resistor is dependent on the off time of the synchroniza-

tion pulse, TOFF(SYNC). Table 1 shows the range of resistors to

be used for a given TOFF(SYNC).

TABLE 1.

TOFF(SYNC) (µsec) RSYNC range (kΩ)

1 5 to 13

2 20 to 40

3 40 to 65

4 55 to 90

5 70 to 110

6 85 to 140

7 100 to 160

8 120 to 190

9 135 to 215

10 150 to 240

It is also necessary to have the width of the synchronization

pulse wider than the duty cycle of the converter (when DR pin

is high and the switching point is low). It is also necessary to

have the synchronization pulse width ≥ 300nsecs.

The FA/SYNC/SD pin also functions as a shutdown pin. If a

high signal (refer to the electrical characteristics for definition

of high signal) appears on the FA/SYNC/SD pin, the LM3488

stops switching and goes into a low current mode. The total

supply current of the IC reduces to less than 10µA under these

conditions.

Figure 8 and Figure 9 show implementation of shutdown func-

tion when operating in Frequency adjust mode and synchro-

nization mode respectively. In frequency adjust mode,

connecting the FA/SYNC/SD pin to ground forces the clock

to run at a certain frequency. Pulling this pin high shuts down

the IC. In frequency adjust or synchronization mode, a high

signal for more than 30µs shuts down the IC.

Figure 10 shows implementation of both frequency adjust with

RFA resistor and frequency synchronization with RSYNC. The

switching frequency is defined by RFA when a synchronization

signal is not applied. When sync is applied it overrides the

RFA setting.

10138814

FIGURE 6. Frequency Adjust

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LM3488/LM3488Q

10138815

FIGURE 7. Frequency Synchronization

10138816

FIGURE 8. Shutdown Operation in Frequency Adjust Mode

10138817

FIGURE 9. Shutdown Operation in Synchronization Mode

10138887

FIGURE 10. Frequency Adjust or Frequency Synchronization

SHORT-CIRCUIT PROTECTION

When the voltage across the sense resistor (measured on

ISEN Pin) exceeds 350mV, short-circuit current limit gets acti-

vated. A comparator inside LM3488 reduces the switching

frequency by a factor of 5 and maintains this condition till the

short is removed.

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LM3488/LM3488Q

Typical Applications

The LM3488 may be operated in either continuous or discon-

tinuous conduction mode. The following applications are de-

signed for continuous conduction operation. This mode of

operation has higher efficiency and lower EMI characteristics

than the discontinuous mode.

BOOST CONVERTER

The most common topology for LM3488 is the boost or step-

up topology. The boost converter converts a low input voltage

into a higher output voltage. The basic configuration for a

boost regulator is shown in Figure 11. In continuous conduc-

tion mode (when the inductor current never reaches zero at

steady state), the boost regulator operates in two cycles. In

the first cycle of operation, MOSFET Q is turned on and en-

ergy is stored in the inductor. During this cycle, diode D is

reverse biased and load current is supplied by the output ca-

pacitor, COUT.

In the second cycle, MOSFET Q is off and the diode is forward

biased. The energy stored in the inductor is transferred to the

load and output capacitor. The ratio of these two cycles de-

termines the output voltage. The output voltage is defined as:

(ignoring the drop across the MOSFET and the diode), or

where D is the duty cycle of the switch, VD is the forward volt-

age drop of the diode, and VQ is the drop across the MOSFET

when it is on. The following sections describe selection of

components for a boost converter.

10138822

FIGURE 11. Simplified Boost Converter Diagram (a) First cycle of operation. (b) Second cycle of operation

POWER INDUCTOR SELECTION

The inductor is one of the two energy storage elements in a

boost converter. Figure 12 shows how the inductor current

varies during a switching cycle. The current through an in-

ductor is quantified as:

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LM3488/LM3488Q

10138824

FIGURE 12. A. Inductor Current B. Diode Current

If VL(t) is constant, diL(t)/dt must be constant. Hence, for a

given input voltage and output voltage, the current in the in-

ductor changes at a constant rate.

The important quantities in determining a proper inductance

value are IL (the average inductor current) and ΔiL (the induc-

tor current ripple). If ΔiL is larger than IL, the inductor current

will drop to zero for a portion of the cycle and the converter

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LM3488/LM3488Q

will operate in discontinuous conduction mode. If ΔiL is small-

er than IL, the inductor current will stay above zero and the

converter will operate in continuous conduction mode. All the

analysis in this datasheet assumes operation in continuous

conduction mode. To operate in continuous conduction

mode, the following conditions must be met:

IL > ΔiL

Choose the minimum IOUT to determine the minimum L. A

common choice is to set ΔiL to 30% of IL. Choosing an ap-

propriate core size for the inductor involves calculating the

average and peak currents expected through the inductor. In

a boost converter,

and IL_peak = IL(max) + ΔiL(max),

where

A core size with ratings higher than these values should be

chosen. If the core is not properly rated, saturation will dra-

matically reduce overall efficiency.

The LM3488 can be set to switch at very high frequencies.

When the switching frequency is high, the converter can be

operated with very small inductor values. With a small induc-

tor value, the peak inductor current can be extremely higher

than the output currents, especially under light load condi-

tions.

The LM3488 senses the peak current through the switch. The

peak current through the switch is the same as the peak cur-

rent calculated above.

PROGRAMMING THE OUTPUT VOLTAGE

The output voltage can be programmed using a resistor di-

vider between the output and the feedback pins, as shown in

Figure 13. The resistors are selected such that the voltage at

the feedback pin is 1.26V. RF1 and RF2 can be selected using

the equation,

A 100pF capacitor may be connected between the feedback

and ground pins to reduce noise.

10138820

FIGURE 13. Adjusting the Output Voltage

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LM3488/LM3488Q

SETTING THE CURRENT LIMIT

The maximum amount of current that can be delivered to the

load is set by the sense resistor, RSEN. Current limit occurs

when the voltage that is generated across the sense resistor

equals the current sense threshold voltage, VSENSE. When

this threshold is reached, the switch will be turned off until the

next cycle. Limits for VSENSE are specified in the electrical

characteristics section. VSENSE represents the maximum val-

ue of the internal control signal VCS. This control signal,

however, is not a constant value and changes over the course

of a period as a result of the internal compensation ramp (See

Figure 3). Therefore the current limit threshold will also

change. The actual current limit threshold is a function of the

sense voltage (VSENSE) and the internal compensation ramp:

RSEN x ISWLIMIT = VCSMAX = VSENSE - (D x VSL)

Where ISWLIMIT is the peak switch current limit, defined by the

equation below. As duty cycle increases, the control voltage

is reduced as VSL ramps up. Since current limit threshold

varies with duty cycle, the following equation should be used

to select RSEN and set the desired current limit threshold:

The numerator of the above equation is VCS, and ISWLIMIT is

calculated as:

To avoid false triggering, the current limit value should have

some margin above the maximum operating value, typically

120%. Values for both VSENSE and VSL are specified in the

characteristic table. However, calculating with the limits of

these two specs could result in an unrealistically wide current

limit or RSEN range. Therefore, the following equation is rec-

ommended, using the VSL ratio value given in the EC table:

RSEN is part of the current mode control loop and has some

influence on control loop stability. Therefore, once the current

limit threshold is set, loop stability must be verified. To verify

stability, use the following equation:

If the selected RSEN is greater than this value, additional slope

compensation must be added to ensure stability, as described

in the section below.

CURRENT LIMIT WITH EXTERNAL SLOPE

COMPENSATION

RSL is used to add additional slope compensation when re-

quired. It is not necessary in most designs and RSL should be

no larger than necessary. Select RSL according to the follow-

ing equation:

Where RSEN is the selected value based on current limit. With

RSL installed, the control signal includes additional external

slope to stabilize the loop, which will also have an effect on

the current limit threshold. Therefore, the current limit thresh-

old must be re-verified, as illustrated in the equations below :

VCS = VSENSE – (D x (VSL + ΔVSL))

Where ΔVSL is the additional slope compensation generated

and calculated as:

ΔVSL = 40 µA x RSL

This changes the equation for current limit (or RSEN) to:

The RSEN and RSL values may have to be calculated iteratively

in order to achieve both the desired current limit and stable

operation. In some designs RSL can also help to filter noise

on the ISEN pin.

If the inductor is selected such that ripple current is the rec-

ommended 30% value, and the current limit threshold is 120%

of the maximum peak, a simpler method can be used to de-

termine RSEN. The equation below will provide optimum sta-

bility without RSL, provided that the above 2 conditions are

met:

POWER DIODE SELECTION

Observation of the boost converter circuit shows that the av-

erage current through the diode is the average load current,

and the peak current through the diode is the peak current

through the inductor. The diode should be rated to handle

more than its peak current. The peak diode current can be

calculated using the formula:

ID(Peak) = IOUT/ (1−D) + ΔIL

In the above equation, IOUT is the output current and ΔIL has

been defined in Figure 12

The peak reverse voltage for boost converter is equal to the

regulator output voltage. The diode must be capable of han-

dling this voltage. To improve efficiency, a low forward drop

schottky diode is recommended.

POWER MOSFET SELECTION

The drive pin of LM3488 must be connected to the gate of an

external MOSFET. In a boost topology, the drain of the ex-

ternal N-Channel MOSFET is connected to the inductor and

the source is connected to the ground. The drive pin (DR)

voltage depends on the input voltage (see typical perfor-

mance characteristics). In most applications, a logic level

MOSFET can be used. For very low input voltages, a sub-

logic level MOSFET should be used.

The selected MOSFET directly controls the efficiency. The

critical parameters for selection of a MOSFET are:

1. Minimum threshold voltage, VTH(MIN)

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LM3488/LM3488Q

2. On-resistance, RDS(ON)

3. Total gate charge, Qg

4. Reverse transfer capacitance, CRSS

5. Maximum drain to source voltage, VDS(MAX)

The off-state voltage of the MOSFET is approximately equal

to the output voltage. VDS(MAX) of the MOSFET must be

greater than the output voltage. The power losses in the

MOSFET can be categorized into conduction losses and ac

switching or transition losses. RDS(ON) is needed to estimate

the conduction losses. The conduction loss, PCOND, is the

I2R loss across the MOSFET. The maximum conduction loss

is given by:

where DMAX is the maximum duty cycle.

The turn-on and turn-off transitions of a MOSFET require

times of tens of nano-seconds. CRSS and Qg are needed to

estimate the large instantaneous power loss that occurs dur-

ing these transitions.

The amount of gate current required to turn the MOSFET on

can be calculated using the formula:

IG = Qg.FS

The required gate drive power to turn the MOSFET on is equal

to the switching frequency times the energy required to deliver

the charge to bring the gate charge voltage to VDR (see elec-

trical characteristics and typical performance characteristics

for the drive voltage specification).

PDrive = FS.Qg.VDR

INPUT CAPACITOR SELECTION

Due to the presence of an inductor at the input of a boost

converter, the input current waveform is continuous and tri-

angular, as shown in Figure 12. The inductor ensures that the

input capacitor sees fairly low ripple currents. However, as the

input capacitor gets smaller, the input ripple goes up. The rms

current in the input capacitor is given by:

The input capacitor should be capable of handling the rms

current. Although the input capacitor is not as critical in a

boost application, low values can cause impedance interac-

tions. Therefore a good quality capacitor should be chosen in

the range of 10µF to 20µF. If a value lower than 10µF is used,

then problems with impedance interactions or switching noise

can affect the LM3478. To improve performance, especially

with VIN below 8 volts, it is recommended to use a 20Ω resistor

at the input to provide a RC filter. The resistor is placed in

series with the VIN pin with only a bypass capacitor attached

to the VIN pin directly (see Figure 14). A 0.1µF or 1µF ceramic

capacitor is necessary in this configuration. The bulk input

capacitor and inductor will connect on the other side of the

resistor with the input power supply.

10138893

FIGURE 14. Reducing IC Input Noise

OUTPUT CAPACITOR SELECTION

The output capacitor in a boost converter provides all the out-

put current when the inductor is charging. As a result it sees

very large ripple currents. The output capacitor should be ca-

pable of handling the maximum rms current. The rms current

in the output capacitor is:

Where

and D, the duty cycle is equal to (VOUT − VIN)/VOUT.

The ESR and ESL of the output capacitor directly control the

output ripple. Use capacitors with low ESR and ESL at the

output for high efficiency and low ripple voltage. Surface

Mount tantalums, surface mount polymer electrolytic and

polymer tantalum, Sanyo- OSCON, or multi-layer ceramic ca-

pacitors are recommended at the output.

Designing SEPIC Using LM3488

Since the LM3488 controls a low-side N-Channel MOSFET,

it can also be used in SEPIC (Single Ended Primary Induc-

tance Converter) applications. An example of SEPIC using

LM3488 is shown in Figure 15. As shown in Figure 15, the

output voltage can be higher or lower than the input voltage.

The SEPIC uses two inductors to step-up or step-down the

input voltage. The inductors L1 and L2 can be two discrete

inductors or two windings of a coupled transformer since

equal voltages are applied across the inductor throughout the

switching cycle. Using two discrete inductors allows use of

catalog magnetics, as opposed to a custom transformer. The

input ripple can be reduced along with size by using the cou-

pled windings of transformer for L1 and L2.

Due to the presence of the inductor L1 at the input, the SEPIC

inherits all the benefits of a boost converter. One main ad-

vantage of SEPIC over boost converter is the inherent input

to output isolation. The capacitor CS isolates the input from

the output and provides protection against shorted or mal-

functioning load. Hence, the A SEPIC is useful for replacing

boost circuits when true shutdown is required. This means

that the output voltage falls to 0V when the switch is turned

off. In a boost converter, the output can only fall to the input

voltage minus a diode drop.

The duty cycle of a SEPIC is given by:

www.national.com 18

LM3488/LM3488Q

In the above equation, VQ is the on-state voltage of the MOS-

FET, Q, and VDIODE is the forward voltage drop of the diode.

10138844

FIGURE 15. Typical SEPIC Converter

POWER MOSFET SELECTION

As in boost converter, the parameters governing the selection

of the MOSFET are the minimum threshold voltage, VTH

(MIN), the on-resistance, RDS(ON), the total gate charge, Qg, the

reverse transfer capacitance, CRSS, and the maximum drain

to source voltage, VDS(MAX). The peak switch voltage in a

SEPIC is given by:

VSW(PEAK) = VIN + VOUT + VDIODE

The selected MOSFET should satisfy the condition:

VDS(MAX) > VSW(PEAK)

The peak switch current is given by:

The rms current through the switch is given by:

POWER DIODE SELECTION

The Power diode must be selected to handle the peak current

and the peak reverse voltage. In a SEPIC, the diode peak

current is the same as the switch peak current. The off-state

voltage or peak reverse voltage of the diode is VIN + VOUT.

Similar to the boost converter, the average diode current is

equal to the output current. Schottky diodes are recommend-

ed.

SELECTION OF INDUCTORS L1 AND L2

Proper selection of the inductors L1 and L2 to maintain con-

stant current mode requires calculations of the following pa-

rameters.

Average current in the inductors:

IL2AVE = IOUT

Peak to peak ripple current, to calculate core loss if neces-

sary:

maintains the condition IL > ΔiL/2 to ensure constant current

mode.

Peak current in the inductor, to ensure the inductor does not

saturate:

19 www.national.com

LM3488/LM3488Q

IL1PK must be lower than the maximum current rating set by

the current sense resistor.

The value of L1 can be increased above the minimum rec-

ommended to reduce input ripple and output ripple. However,

once DIL1 is less than 20% of IL1AVE, the benefit to output ripple

is minimal.

By increasing the value of L2 above the minimum recom-

mended, ΔIL2 can be reduced, which in turn will reduce the

output ripple voltage:

where ESR is the effective series resistance of the output ca-

pacitor.

If L1 and L2 are wound on the same core, then L1 = L2 = L.

All the equations above will hold true if the inductance is re-

placed by 2L. A good choice for transformer with equal turns

is Coiltronics CTX series Octopack.

SENSE RESISTOR SELECTION

The peak current through the switch, ISW(PEAK) can be adjust-

ed using the current sense resistor, RSEN, to provide a certain

output current. Resistor RSEN can be selected using the for-

mula:

Sepic Capacitor Selection

The selection of SEPIC capacitor, CS, depends on the rms

current. The rms current of the SEPIC capacitor is given by:

The SEPIC capacitor must be rated for a large ACrms current

relative to the output power. This property makes the SEPIC

much better suited to lower power applications where the rms

current through the capacitor is relatively small (relative to

capacitor technology). The voltage rating of the SEPIC ca-

pacitor must be greater than the maximum input voltage.

Tantalum capacitors are the best choice for SMT, having high

rms current ratings relative to size. Ceramic capacitors could

be used, but the low C values will tend to cause larger

changes in voltage across the capacitor due to the large cur-

rents. High C value ceramics are expensive. Electrolytics

work well for through hole applications where the size re-

quired to meet the rms current rating can be accommodated.

There is an energy balance between CS and L1, which can

be used to determine the value of the capacitor. The basic

energy balance equation is:

Where

is the ripple voltage across the SEPIC capacitor, and

is the ripple current through the inductor L1. The energy bal-

ance equation can be solved to provide a minimum value for

CS:

Input Capacitor Selection

Similar to a boost converter, the SEPIC has an inductor at the

input. Hence, the input current waveform is continuous and

triangular. The inductor ensures that the input capacitor sees

fairly low ripple currents. However, as the input capacitor gets

smaller, the input ripple goes up. The rms current in the input

capacitor is given by:

The input capacitor should be capable of handling the rms

current. Although the input capacitor is not as critical in a

boost application, low values can cause impedance interac-

tions. Therefore a good quality capacitor should be chosen in

the range of 10µF to 20µF. If a value lower than 10µF is used,

then problems with impedance interactions or switching noise

can affect the LM3478. To improve performance, especially

with VIN below 8 volts, it is recommended to use a 20Ω resistor

at the input to provide a RC filter. The resistor is placed in

series with the VIN pin with only a bypass capacitor attached

to the VIN pin directly (see Figure 14). A 0.1µF or 1µF ceramic

capacitor is necessary in this configuration. The bulk input

capacitor and inductor will connect on the other side of the

resistor with the input power supply.

www.national.com 20

LM3488/LM3488Q

Output Capacitor Selection

The ESR and ESL of the output capacitor directly control the

output ripple. Use low capacitors with low ESR and ESL at

the output for high efficiency and low ripple voltage. Surface

mount tantalums, surface mount polymer electrolytic and

polymer tantalum, Sanyo- OSCON, or multi-layer ceramic ca-

pacitors are recommended at the output.

The output capacitor of the SEPIC sees very large ripple cur-

rents (similar to the output capacitor of a boost converter. The

rms current through the output capacitor is given by:

The ESR and ESL of the output capacitor directly control the

output ripple. Use low capacitors with low ESR and ESL at

the output for high efficiency and low ripple voltage. Surface

mount tantalums, surface mount polymer electrolytic and

polymer tantalum, Sanyo- OSCON, or multi-layer ceramic ca-

pacitors are recommended at the output for low ripple.

Other Application Circuits

10138843

FIGURE 16. Typical High Efficiency Step-Up (Boost) Converter

21 www.national.com

LM3488/LM3488Q

Physical Dimensions inches (millimeters) unless otherwise noted

www.national.com 22

LM3488/LM3488Q

Notes

23 www.national.com

LM3488/LM3488Q

Notes

LM3488/LM3488Q High Efficiency Low-Side N-Channel Controller for Switching Regulators

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