REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
a
ADP1111
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700 World Wide Web Site: http://www.analog.com
Fax: 617/326-8703 © Analog Devices, Inc., 1996
Micropower, Step-Up/Step-Down SW
Regulator; Adjustable and Fixed 3.3 V, 5 V, 12 V
FUNCTIONAL BLOCK DIAGRAMS
DRIVER
ILIM
SW1
SW2
VIN
GND
SET
A0
GAIN BLOCK/
ERROR AMP
COMPARATOR
A1
A2
FB
1.25V
REFERENCE
OSCILLATOR
ADP1111
DRIVER
ILIM
SW1
SW2
VIN
GND
SET
A0
GAIN BLOCK/
ERROR AMP
COMPARATOR
A1
A2
SENSE
1.25V
REFERENCE
OSCILLATOR
ADP1111-5
ADP1111-12
R1 R2 220k
FEATURES
Operates from 2 V to 30 V Input Voltage Range
72 kHz Frequency Operation
Utilizes Surface Mount Inductors
Very Few External Components Required
Operates in Step-Up/Step-Down or Inverting Mode
Low Battery Detector
User Adjustable Current Limit
Internal 1 A Power Switch
Fixed or Adjustable Output Voltage
8-Pin DIP or SO-8 Package
APPLICATIONS
3 V to 5 V, 5 V to 12 V Step-Up Converters
9 V to 5 V, 12 V to 5 V Step-Down Converters
Laptop and Palmtop Computers
Cellular Telephones
Flash Memory VPP Generators
Remote Controls
Peripherals and Add-On Cards
Battery Backup Supplies
Uninterruptible Supplies
Portable Instruments
GENERAL DESCRIPTION
The ADP1111 is part of a family of step-up/step-down switch-
ing regulators that operates from an input voltage supply of 2 V
to 12 V in step-up mode and up to 30 V in step-down mode.
The ADP1111 can be programmed to operate in step-up/step-
down or inverting applications with only 3 external components.
The fixed outputs are 3.3 V, 5 V and 12 V; and an adjustable
version is also available. The ADP1111 can deliver 100 mA at
5 V from a 3 V input in step-up mode, or it can deliver 200 mA
at 5 V from a 12 V input in step-down mode.
Maximum switch current can be programmed with a single
resistor, and an open collector gain block can be arranged in
multiple configuration for low battery detection, as a post linear
regulator, undervoltage lockout, or as an error amplifier.
If input voltages are lower than 2 V, see the ADP1110.
REV. A
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700 www.analog.com
Fax: 781.461.3113 ©2009 Analog Devices, Inc. All rights reserved.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700 www.analog.com
Fax: 781.461.3113 ©1996–2009 Analog Devices, Inc. All rights reserved.
–2– REV. 0
ADP1111–SPECIFICATIONS
Parameter Conditions V
S
Min Typ Max Units
QUIESCENT CURRENT Switch Off I
Q
300 500 μA
INPUT VOLTAGE Step-Up Mode V
IN
2.0 12.6 V
Step-Down Mode 30.0 V
COMPARATOR TRIP POINT
VOLTAGE ADP1111
1
1.20 1.25 1.30 V
OUTPUT SENSE VOLTAGE ADP1111-3.3 V
OUT
3.13 3.30 3.47 V
ADP1111-5
2
4.75 5.00 5.25 V
ADP1111-12
2
11.40 12.00 12.60 V
COMPARATOR HYSTERESIS ADP1111 8 12.5 mV
OUTPUT HYSTERESIS ADP1111-3.3 21 50 mV
ADP1111-5 32 50 mV
ADP1111-12 75 120 mV
OSCILLATOR FREQUENCY f
OSC
54 72 88 kHz
DUTY CYCLE Full Load DC 43 50 65 %
SWITCH ON TIME I
LIM
Tied to V
IN
t
ON
57 9 μs
SW SATURATION VOLTAGE T
A
= +25°C
STEP-UP MODE V
IN
= 3.0 V, I
SW
= 650 mA V
SAT
0.5 0.65 V
V
IN
= 5.0 V, I
SW
= 1 A 0.8 1.0 V
STEP-DOWN MODE V
IN
= 12 V, I
SW
= 650 mA 1.1 1.5 V
FEEDBACK PIN BIAS CURRENT ADP1111 V
FB
= 0 V I
FB
160 300 nA
SET PIN BIAS CURRENT V
SET
= V
REF
I
SET
270 400 nA
GAIN BLOCK OUTPUT LOW I
SINK
= 300 μA
V
SET
= 1.00 V V
OL
0.15 0.4 V
REFERENCE LINE REGULATION 5 V ≤ V
IN
≤ 30 V 0.02 0.075 %/V
2 V ≤ V
IN
≤ 5 V 0.4 %/V
GAIN BLOCK GAIN R
L
= 100 kΩ
3
A
V
1000 6000 V/V
CURRENT LIMIT T
A
= +25°C
220 Ω from I
LIM
to V
IN
I
LIM
400 mA
CURRENT LIMIT TEMPERATURE
COEFFICIENT –0.3 %/°C
SWITCH OFF LEAKAGE CURRENT T
A
= +25°C
Measured at SW1 Pin
V
SW1
= 12 V 1 10 μA
MAXIMUM EXCURSION BELOW GND T
A
= +25°C
I
SW1
≤ 10 μA, Switch Off –400 –350 mV
NOTES
1
This specification guarantees that both the high and low trip points of the comparator fall within the 1.20 V to 1.30 V range.
2
The output voltage waveform will exhibit a sawtooth shape due to the comparator hysteresis. The output voltage on the fixed output versions will always be within
the specified range.
3
100 kΩ resistor connected between a 5 V source and the AO pin.
4
All limits at temperature extremes are guaranteed via correlation using standard statistical methods.
Specifications subject to change without notice.
(0ⴗC ≤ TA ≤ +70ⴗC, VIN = 3 V unless otherwise noted)
REV. A
ADP1111
REV.A 3
ABSOLUTE MAXIMUM RATINGS
Parameter Rating
Supply Voltage 36 V
SW1 Pin Voltage 50 V
SW2 Pin Voltage −0.5 V to VIN
Feedback Pin Voltage (ADP1111) 5.5 V
Switch Current 1.5 A
Maximum Power Dissipation 500 mW
Operating Temperature Range
ADP1111A 0°C to 70°C
Storage Temperature Range −65°C to +150°C
Lead Temperature (Soldering, 10 sec) 300°C
TYPICAL APPLICATION
10µF
(OPTIONAL)
I
LIM
V
IN
SW1
SW2GND
ADP1111AR-5
SUMID
A
CD54-220K
22µH
5V
100mA
33µF
SENSE
3V
INPUT
MBRS120T3
Figure 1. 3 V to 5 V Step-Up Converter
ESD CAUTION
PIN DESCRIPTIONS
Mnemonic Function
ILIM For normal conditions this pin is connected to VIN.
When lower current is required, a resistor should be
connected between ILIM and VIN. Limiting the switch
current to 400 mA is achieved by connecting a 220 Ω
resistor.
VIN Input Voltage.
SW1 Collector Node of Power Transistor. For step-down
con-figuration, connect to VIN. For step-up configu-
ration, connect to an inductor/diode.
SW2 Emitter Node of Power Transistor. For step-down
configuration, connect to inductor/diode. For step-up
configuration, connect to ground. Do not allow this
pin to go more than a diode drop below ground.
GND Ground.
AO Auxiliary Gain (GB) Output. The open collector can
sink 300 μA. It can be left open if unused.
SET Gain Amplifier Input. The amplifier’s positive input is
connected to SET pin and its negative input is con-
nected to the 1.25 V reference. It can be left open if
unused.
FB/SENSE On the ADP1111 (adjustable) version this pin is con-
nected to the comparator input. On the ADP1111-3.3,
ADP1111-5 and ADP1111-12, the pin goes directly
to the internal application resistor that sets output
voltage.
PIN CONFIGURATIONS
A0
I
LIM
S
W1
GND
V
IN
S
W2
*FIXED VERSIONS
FB (SENSE)*
SET
1
2
3
4
8
7
6
5
ADP1111
TOP VIEW
(Not to Scale)
8-Lead Plastic DIP
(N-8)
A0
I
LIM
SW1
GND
V
IN
SW2
*FIXED VERSIONS
FB (SENSE)
*
SET
1
2
3
4
8
7
6
5
ADP1111
TOP VIEW
(Not to Scale)
8-Lead Plastic SOIC
(SO-8)
ISWITCH CURRENT – A
1.4
0
0.1 0.2 0.4 0.6 0.8 1.0 1.2
1.2
0.6
0.4
0.2
1.0
0.8
VIN = 2V
VIN = 3V
VIN = 5V
SATURATION VOLTAGE – V
Figure 2. Saturation Voltage vs. I
SWITCH
Current in
Step-Up Mode
I
SWITCH
CURRENT – A
2.0
0
0.1 0.2 0.4 0.6 0.8 0.9
1.8
0.6
0.4
0.2
1.6
1.4
V
IN
= 12V
ON VOLTAGE – V
1.2
1.0
0.8
Figure 3. Switch ON Voltage vs. I
SWITCH
Current In
Step-Down Mode
INPUT VOLTAGE – V
1.5 303 6 9 12 15 18 21 24 27
0
1400
1200
1000
800
600
400
200
QUIESCENT CURRENT
QUIESCENT CURRENT – μA
Figure 4. Quiescent Current vs. Input Voltage
ADP1111–Typical Characteristics
–4– REV. 0
76
71
67
2304 6 8 1012151821 2427
75
72
70
69
74
73
INPUT VOLTAGE – V
OSCILLATOR FREQUENCY – kHz
OSCILLATOR FREQUENCY
68
Figure 5. Oscillator Frequency vs. Input Voltage
R
LIM
– Ω
1.9
1.7
0.1
1 100010 100
1.5
1.3
0.5
1.1
0.9
0.7
0.3
SWITCH CURRENT – A
STEP-DOWN WITH
V
IN
= 12V
STEP-UP WITH
2V < V
IN
< 5V
Figure 6. Maximum Switch Current vs. R
LIM
OSCILLATOR FREQUENCY – kHz
TEMPERATURE – ⴗC
80
70
60
–40 8525
64
62
68
66
OSCILLATOR FREQUENCY
70
0
78
72
76
74
Figure 7. Oscillator Frequency vs. Temperature
REV. A
ADP1111
–5–
REV. 0
TEMPERATURE – ⴗC
7.5
6.6
–40 8525
7.3
7.2
6.8
6.7
7.4
ON TIME
ON TIME – μs
7.1
6.9
7.0
070
Figure 8. Switch ON Time vs. Temperature
TEMPERATURE – ⴗC
58
56
46
–40 8525
54
48
DUTY CYCLE
DUTY CYCLE – %
52
50
070
Figure 9. Duty Cycle vs. Temperature
TEMPERATURE – ⴗC
0.6
0.5
0
–40 8525
0.3
0.2
0.1
0.4
V
IN
= 3V
@
I
SW
= 0.65A
SATURATION VOLTAGE – V
070
Figure 10. Saturation Voltage vs. Temperature in Step-Up
Mode
TEMPERATURE – ⴗC
1.10
1.05
0.80
–40 8525
1.00
0.85
V
IN
= 12V
@
I
SW
= 0.65A
ON VOLTAGE – V
0.95
0.90
070
Figure 11. Switch ON Voltage vs. Temperature in Step-
Down Mode
TEMPERATURE – ⴗC
500
0
–40 8525
300
250
200
100
400
350 QUIESCENT CURRENT
QUIESCENT CURRENT – μA
150
50
070
450
Figure 12. Quiescent Current vs. Temperature
TEMPERATURE – ⴗC
250
0
–40 8525
150
100
50
200
BIAS CURRENT
BIAS CURRENT – μA
070
Figure 13. Feedback Bias Current vs. Temperature
REV. A
ADP1111
–6– REV. 0
THEORY OF OPERATION
The ADP1111 is a flexible, low-power, switch-mode power
supply (SMPS) controller. The regulated output voltage can be
greater than the input voltage (boost or step-up mode) or less
than the input (buck or step-down mode). This device uses a
gated-oscillator technique to provide very high performance
with low quiescent current.
A functional block diagram of the ADP1111 is shown on
the first page of this data sheet. The internal 1.25 V reference is
connected to one input of the comparator, while the other input
is externally connected (via the FB pin) to a feedback network
connected to the regulated output. When the voltage at the FB
pin falls below 1.25 V, the 72 kHz oscillator turns on. A driver
amplifier provides base drive to the internal power switch, and
the switching action raises the output voltage. When the voltage
at the FB pin exceeds 1.25 V, the oscillator is shut off. While
the oscillator is off, the ADP1111 quiescent current is only
300 μA. The comparator includes a small amount of hysteresis,
which ensures loop stability without requiring external compo-
nents for frequency compensation.
The maximum current in the internal power switch can be set
by connecting a resistor between V
IN
and the I
LIM
pin. When the
maximum current is exceeded, the switch is turned OFF. The
current limit circuitry has a time delay of about 1 μs. If an
external resistor is not used, connect I
LIM
to V
IN
. Further
information on I
LIM
is included in the “APPLICATIONS”
section of this data sheet.
The ADP1111 internal oscillator provides 7 μs ON and 7 μs
OFF times that are ideal for applications where the ratio
between V
IN
and V
OUT
is roughly a factor of two (such as
converting +3 V to + 5 V). However, wider range conversions
(such as generating +12 V from a +5 V supply) can easily be
accomplished.
An uncommitted gain block on the ADP1111 can be connected
as a low-battery detector. The inverting input of the gain block
is internally connected to the 1.25 V reference. The noninverting
input is available at the SET pin. A resistor divider, connected
between V
IN
and GND with the junction connected to the SET
pin, causes the AO output to go LOW when the low battery set
point is exceeded. The AO output is an open collector NPN
transistor that can sink 300 μA.
The ADP1111 provides external connections for both the
collector and emitter of its internal power switch that permit
both step-up and step-down modes of operation. For the step-
up mode, the emitter (Pin SW2) is connected to GND, and the
collector (Pin SW1) drives the inductor. For step-down mode,
the emitter drives the inductor while the collector is connected
to V
IN
.
The output voltage of the ADP1111 is set with two external
resistors. Three fixed-voltage models are also available:
ADP1111–3.3 (+3.3 V), ADP1111–5 (+5 V) and ADP1111–12
(+12 V). The fixed-voltage models are identical to the
ADP1111, except that laser-trimmed voltage-setting resistors
are included on the chip. On the fixed-voltage models of the
ADP1111, simply connect the feedback pin (Pin 8) directly to
the output voltage.
COMPONENT SELECTION
General Notes on Inductor Selection
When the ADP1111 internal power switch turns on, current
begins to flow in the inductor. Energy is stored in the inductor
core while the switch is on, and this stored energy is transferred
to the load when the switch turns off. Since both the collector
and the emitter of the switch transistor are accessible on the
ADP1111, the output voltage can be higher, lower, or of
opposite polarity than the input voltage.
To specify an inductor for the ADP1111, the proper values of
inductance, saturation current and dc resistance must be
determined. This process is not difficult, and specific equations
for each circuit configuration are provided in this data sheet. In
general terms, however, the inductance value must be low
enough to store the required amount of energy (when both
input voltage and switch ON time are at a minimum) but high
enough that the inductor will not saturate when both V
IN
and
switch ON time are at their maximum values. The inductor
must also store enough energy to supply the load, without
saturating. Finally, the dc resistance of the inductor should be
low so that excessive power will not be wasted by heating the
windings. For most ADP1111 applications, an inductor of
15 μH to 100 μH with a saturation current rating of 300 mA to
1 A and dc resistance <0.4 Ω is suitable. Ferrite-core inductors
that meet these specifications are available in small, surface-
mount packages.
To minimize Electro-Magnetic Interference (EMI), a toroid or
pot-core type inductor is recommended. Rod-core inductors are
a lower-cost alternative if EMI is not a problem.
CALCULATING THE INDUCTOR VALUE
Selecting the proper inductor value is a simple three step
process:
1. Define the operating parameters: minimum input voltage,
maximum input voltage, output voltage and output current.
2. Select the appropriate conversion topology (step-up, step-
down, or inverting).
3. Calculate the inductor value using the equations in the
following sections.
TEMPERATURE – ⴗC
350
300
0
–40 8525
200
150
100
50
250 BIAS CURRENT
BIAS CURRENT – μA
070
Figure 14. Set Pin Bias Current vs. Temperature
REV. A
ADP1111
–7–
REV. 0
INDUCTOR SELECTION–STEP-UP CONVERTER
In a step-up or boost converter (Figure 18), the inductor must
store enough power to make up the difference between the input
voltage and the output voltage. The power that must be stored
is calculated from the equation:
P
L
=V
OUT
+V
D
−V
IN(MIN )
()
•I
OUT
()
(Equation 1)
where V
D
is the diode forward voltage (0.5 V for a 1N5818
Schottky). Because energy is only stored in the inductor while
the ADP1111 switch is ON, the energy stored in the inductor
on each switching cycle must be equal to or greater than:
P
f
L
OSC
(Equation 2)
in order for the ADP1111 to regulate the output voltage.
When the internal power switch turns ON, current flow in the
inductor increases at the rate of:
I
L
t
()
=V
IN
R'1−e
−R't
L
⎛
⎝
⎜⎞
⎠
⎟
(Equation 3)
where L is in Henrys and R' is the sum of the switch equivalent
resistance (typically 0.8 Ω at +25°C) and the dc resistance of
the inductor. In most applications, the voltage drop across the
switch is small compared to V
IN
so a simpler equation can be
used:
I
L
t
()
=V
IN
Lt
(Equation 4)
Replacing ‘t’ in the above equation with the ON time of the
ADP1111 (7 μs, typical) will define the peak current for a given
inductor value and input voltage. At this point, the inductor
energy can be calculated as follows:
E
L
=1
2L•I
2
PEAK
(Equation 5)
As previously mentioned, E
L
must be greater than P
L
/f
OSC
so
that the ADP1111 can deliver the necessary power to the load.
For best efficiency, peak current should be limited to 1 A or
less. Higher switch currents will reduce efficiency because of
increased saturation voltage in the switch. High peak current
also increases output ripple. As a general rule, keep peak current
as low as possible to minimize losses in the switch, inductor and
diode.
In practice, the inductor value is easily selected using the
equations above. For example, consider a supply that will
generate 12 V at 40 mA from a 9 V battery, assuming a 6 V
end-of-life voltage. The inductor power required is, from
Equation 1:
P
L
=12V+0.5V−6V
()
•40 mA
()
=260 mW
On each switching cycle, the inductor must supply:
P
L
f
OSC
=260 mW
72 kHz =3.6 μJ
Since the required inductor power is fairly low in this example,
the peak current can also be low. Assuming a peak current of
500 mA as a starting point, Equation 4 can be rearranged to
recommend an inductor value:
L=V
IN
I
L(MAX )
t=6V
500 mA 7μs=84 μH
Substituting a standard inductor value of 68 μH with 0.2 Ω dc
resistance will produce a peak switch current of:
I
PEAK
=6V
1. 0 Ω1−e
−1.0 Ω•7μs
68 μH
⎛
⎝
⎜⎞
⎠
⎟=587 mA
Once the peak current is known, the inductor energy can be
calculated from Equation 5:
EL=1
268 μH
()
•587 mA
()
2=11.7 μJ
Since the inductor energy of 11.7 μJ is greater than the P
L
/f
OSC
requirement of 3.6 μJ, the 68 μH inductor will work in this
application. By substituting other inductor values into the same
equations, the optimum inductor value can be selected.
When selecting an inductor, the peak current must not exceed
the maximum switch current of 1.5 A. If the equations shown
above result in peak currents > 1.5 A, the ADP1110 should be
considered. Since this device has a 70% duty cycle, more energy
is stored in the inductor on each cycle. This results is greater
output power.
The peak current must be evaluated for both minimum and
maximum values of input voltage. If the switch current is high
when V
IN
is at its minimum, the 1.5 A limit may be exceeded at
the maximum value of V
IN
. In this case, the ADP1111’s current
limit feature can be used to limit switch current. Simply select a
resistor (using Figure 6) that will limit the maximum switch
current to the I
PEAK
value calculated for the minimum value of
V
IN
. This will improve efficiency by producing a constant I
PEAK
as V
IN
increases. See the “Limiting the Switch Current” section
of this data sheet for more information.
Note that the switch current limit feature does not protect the
circuit if the output is shorted to ground. In this case, current is
only limited by the dc resistance of the inductor and the forward
voltage of the diode.
INDUCTOR SELECTION–STEP-DOWN CONVERTER
The step-down mode of operation is shown in Figure 19.
Unlike the step-up mode, the ADP1111’s power switch does not
saturate when operating in the step-down mode; therefore,
switch current should be limited to 650 mA in this mode. If the
input voltage will vary over a wide range, the I
LIM
pin can be
used to limit the maximum switch current. Higher switch
current is possible by adding an external switching transistor as
shown in Figure 21.
The first step in selecting the step-down inductor is to calculate
the peak switch current as follows:
IPEAK =2IOUT
DC
V
OUT +VD
VIN −VSW +VD
⎛
⎝
⎜⎞
⎠
⎟
(Equation 6)
where DC = duty cycle (0.5 for the ADP1111)
V
SW
= voltage drop across the switch
V
D
= diode drop (0.5 V for a 1N5818)
I
OUT
= output current
V
OUT
= the output voltage
V
IN
= the minimum input voltage
REV. A
ADP1111
–8– REV. 0
As previously mentioned, the switch voltage is higher in step-
down mode than in step-up mode. V
SW
is a function of switch
current and is therefore a function of V
IN
, L, time and V
OUT
.
For most applications, a V
SW
value of 1.5 V is recommended.
The inductor value can now be calculated:
L=V
IN MIN
()
−V
SW
−V
OUT
I
PEAK
•t
ON
(Equation 7)
where t
ON
= switch ON time (7 μs).
If the input voltage will vary (such as an application that must
operate from a 9 V, 12 V or 15 V source), an R
LIM
resistor
should be selected from Figure 6. The R
LIM
resistor will keep
switch current constant as the input voltage rises. Note that
there are separate R
LIM
values for step-up and step-down modes
of operation.
For example, assume that +5 V at 300 mA is required from a
+12 V to +24 V source. Deriving the peak current from
Equation 6 yields:
I
PEAK
=2•300 mA
0.5
5+0.5
12 −1. 5 +0.5
⎛
⎝
⎜⎞
⎠
⎟=600 mA
Then, the peak current can be inserted into Equation 7 to
calculate the inductor value:
L=12 −1. 5 −5
600 mA •7μs=64 μH
Since 64 μH is not a standard value, the next lower standard
value of 56 μH would be specified.
To avoid exceeding the maximum switch current when the
input voltage is at +24 V, an R
LIM
resistor should be specified.
Using the step-down curve of Figure 6, a value of 560 Ω will
limit the switch current to 600 mA.
INDUCTOR SELECTION–POSITIVE-TO-NEGATIVE
CONVERTER
The configuration for a positive-to-negative converter using the
ADP1111 is shown in Figure 22. As with the step-up converter,
all of the output power for the inverting circuit must be supplied
by the inductor. The required inductor power is derived from
the formula:
P = I
L OUT
VV
OUT D
+
()
•
()
(Equation 8)
The ADP1111 power switch does not saturate in positive-to-
negative mode. The voltage drop across the switch can be
modeled as a 0.75 V base-emitter diode in series with a 0.65 Ω
resistor. When the switch turns on, inductor current will rise at
a rate determined by:
I
L
t
()
=V
L
R'1−e
−R't
L
⎛
⎝
⎜⎞
⎠
⎟
(Equation 9)
where: R' = 0.65 Ω + R
L(DC)
V
L
= V
IN
– 0.75 V
For example, assume that a –5 V output at 50 mA is to be
generated from a +4.5 V to +5.5 V source. The power in the
inductor is calculated from Equation 8:
P
L
=|−5V|+0.5V|
()
•50 mA
()
=275 mW
During each switching cycle, the inductor must supply the
following energy:
P
L
f
OSC
=275 mW
72 kHz =3.8 μJ
Using a standard inductor value of 56 μH with 0.2 Ω dc
resistance will produce a peak switch current of:
I
PEAK
=4.5V−0.75V
0.65 Ω+0.2 Ω1−e
−0.85 Ω•7μs
56 μH
⎛
⎝
⎜⎞
⎠
⎟=445 mA
Once the peak current is known, the inductor energy can be
calculated from (Equation 9):
E
L
=1
256 μH
()
•445 mA
()
2
=5.54 μJ
Since the inductor energy of 5.54
μ
J is greater than the P
L
/f
OSC
requirement of 3.82 μJ, the 56 μH inductor will work in this
application.
The input voltage only varies between 4.5 V and 5.5 V in this
application. Therefore, the peak current will not change enough
to require an R
LIM
resistor and the I
LIM
pin can be connected
directly to V
IN
. Care should be taken, of course, to ensure that
the peak current does not exceed 650 mA.
CAPACITOR SELECTION
For optimum performance, the ADP1111’s output capacitor
must be selected carefully. Choosing an inappropriate capacitor
can result in low efficiency and/or high output ripple.
Ordinary aluminum electrolytic capacitors are inexpensive but
often have poor Equivalent Series Resistance (ESR) and
Equivalent Series Inductance (ESL). Low ESR aluminum
capacitors, specifically designed for switch mode converter
applications, are also available, and these are a better choice
than general purpose devices. Even better performance can be
achieved with tantalum capacitors, although their cost is higher.
Very low values of ESR can be achieved by using OS-CON
capacitors (Sanyo Corporation, San Diego, CA). These devices
are fairly small, available with tape-and-reel packaging and have
very low ESR.
The effects of capacitor selection on output ripple are demon-
strated in Figures 15, 16 and 17. These figures show the output
of the same ADP1111 converter that was evaluated with three
different output capacitors. In each case, the peak switch
current is 500 mA, and the capacitor value is 100 μF. Figure 15
shows a Panasonic HF-series 16-volt radial cap. When the
switch turns off, the output voltage jumps by about 90 mV and
then decays as the inductor discharges into the capacitor. The
rise in voltage indicates an ESR of about 0.18 Ω. In Figure 16,
the aluminum electrolytic has been replaced by a Sprague 293D
series, a 6 V tantalum device. In this case the output jumps
about 30 mV, which indicates an ESR of 0.06 Ω. Figure 17
shows an OS-CON 16–volt capacitor in the same circuit, and
ESR is only 0.02 Ω.
REV. A
ADP1111
–9–
REV. 0
Figure 15. Aluminum Electrolytic
Figure 16. Tantalum Electrolytic
Figure 17. OS-CON Capacitor
If low output ripple is important, the user should consider the
ADP3000. Because this device switches at 400 kHz, lower peak
current can be used. Also, the higher switching frequency
simplifies the design of the output filter. Consult the ADP3000
data sheet for additional details.
DIODE SELECTION
In specifying a diode, consideration must be given to speed,
forward voltage drop and reverse leakage current. When the
ADP1111 switch turns off, the diode must turn on rapidly if
high efficiency is to be maintained. Shottky rectifiers, as well as
fast signal diodes such as the 1N4148, are appropriate. The
forward voltage of the diode represents power that is not
delivered to the load, so V
F
must also be minimized. Again,
Schottky diodes are recommended. Leakage current is especially
important in low-current applications where the leakage can be
a significant percentage of the total quiescent current.
For most circuits, the 1N5818 is a suitable companion to the
ADP1111. This diode has a V
F
of 0.5 V at 1 A, 4 μA to 10 μA
leakage, and fast turn-on and turn-off times. A surface mount
version, the MBRS130T3, is also available.
For switch currents of 100 mA or less, a Shottky diode such as
the BAT85 provides a V
F
of 0.8 V at 100 mA and leakage less
than 1 μA. A similar device, the BAT54, is available in a SOT23
package. Even lower leakage, in the 1 nA to 5 nA range, can be
obtained with a 1N4148 signal diode.
General purpose rectifiers, such as the 1N4001, are not suitable
for ADP1111 circuits. These devices, which have turn-on times
of 10 μs or more, are far too slow for switching power supply
applications. Using such a diode “just to get started” will result
in wasted time and effort. Even if an ADP1111 circuit appears
to function with a 1N4001, the resulting performance will not
be indicative of the circuit performance when the correct diode
is used.
CIRCUIT OPERATION, STEP-UP (BOOST) MODE
In boost mode, the ADP1111 produces an output voltage that is
higher than the input voltage. For example, +12 V can be gener-
ated from a +5 V logic power supply or +5 V can be derived
from two alkaline cells (+3 V).
Figure 18 shows an ADP1111 configured for step-up operation.
The collector of the internal power switch is connected to the
output side of the inductor, while the emitter is connected to
GND. When the switch turns on, pin SW1 is pulled near
ground. This action forces a voltage across L1 equal to
V
IN
– V
CE(SAT)
, and current begins to flow through L1. This
current reaches a final value (ignoring second-order effects) of:
I
PEAK
≅V
IN
−V
CE (SAT )
L•7μs
where 7
μ
s is the ADP1111 switch’s “on” time.
I
LIM
V
IN
SW1
FB
GND SW2
ADP1111
5 4
+
V
IN
L1
D1
1N5818
C1
R2
R1
V
OUT
R3
(OPTIONAL)
12
3
8
Figure 18. Step-Up Mode Operation
When the switch turns off, the magnetic field collapses. The
polarity across the inductor changes, current begins to flow
through D1 into the load, and the output voltage is driven above
the input voltage.
The output voltage is fed back to the ADP1111 via resistors R1
and R2. When the voltage at pin FB falls below 1.25 V, SW1
turns “on” again, and the cycle repeats. The output voltage is
therefore set by the formula:
V
OUT
=1. 25 V•1+R2
R1
⎛
⎝
⎜⎞
⎠
⎟
The circuit of Figure 18 shows a direct current path from V
IN
to
V
OUT
, via the inductor and D1. Therefore, the boost converter
is not protected if the output is short circuited to ground.
REV. A
ADP1111
–10– REV. 0
CIRCUIT OPERATION, STEP DOWN (BUCK) MODE)
The ADP1111’s step down mode is used to produce an output
voltage that is lower than the input voltage. For example, the
output of four NiCd cells (+4.8 V) can be converted to a +3 V
logic supply.
A typical configuration for step down operation of the ADP1111
is shown in Figure 19. In this case, the collector of the internal
power switch is connected to V
IN
and the emitter drives the
inductor. When the switch turns on, SW2 is pulled up towards
V
IN
. This forces a voltage across L1 equal to V
IN
– V
CE
– V
OUT
and causes current to flow in L1. This current reaches a final
value of:
I
PEAK
≅V
IN
−V
CE
−V
OUT
L•7μs
where 7
μ
s is the ADP1111 switch’s “on” time.
I
LIM
V
IN
SW1
SW2 4
GNDSETAO
ADP1111
NC
L1
D1
1N5818
R
LIM
100Ω
1
+
V
IN
2 3
67 5
NC
C
2
+
V
OUT
R2
R1
C
L
FB 8
Figure 19. Step-Down Mode Operation
When the switch turns off, the magnetic field collapses. The
polarity across the inductor changes, and the switch side of the
inductor is driven below ground. Schottky diode D1 then turns
on, and current flows into the load. Notice that the Absolute
Maximum Rating for the ADP1111’s SW2 pin is 0.5 V below
ground. To avoid exceeding this limit, D1 must be a Schottky
diode. If a silicon diode is used for D1, Pin SW2 can go to
–0.8 V, which will cause potentially damaging power dissipation
within the ADP1111.
The output voltage of the buck regulator is fed back to the
ADP1111’s FB pin by resistors R1 and R2. When the voltage at
pin FB falls below 1.25 V, the internal power switch turns “on”
again, and the cycle repeats. The output voltage is set by the
formula:
V
OUT
=1. 25 V•1+R2
R1
⎛
⎝
⎜⎞
⎠
⎟
When operating the ADP1111 in step-down mode, the output
voltage is impressed across the internal power switch’s emitter-
base junction when the switch is off. To protect the switch, the
output voltage should be limited to 6.2 V or less. If a higher
output voltage is required, a Schottky diode should be placed in
series with SW2 as shown in Figure 20.
I
LIM
V
IN
SW1
SW2
FB
GND
ADP1111
L1
D1
R
3
1
+
V
IN
2 3
5
8
4
C
2
+
V
OUT
R2
R1
D2
C
1
D1, D2 = 1N5818 SCHOTTKY DIODES
Figure 20. Step-Down Model, V
OUT
> 6.2 V
If the input voltage to the ADP1111 varies over a wide range, a
current limiting resistor at Pin 1 may be required. If a particular
circuit requires high peak inductor current with minimum input
supply voltage, the peak current may exceed the switch maxi-
mum rating and/or saturate the inductor when the supply
voltage is at the maximum value. See the “Limiting the Switch
Current” section of this data sheet for specific recommendations.
INCREASING OUTPUT CURRENT IN THE STEP-DOWN
REGULATOR
Unlike the boost configuration, the ADP1111’s internal power
switch is not saturated when operating in step-down mode. A
conservative value for the voltage across the switch in step-down
mode is 1.5 V. This results in high power dissipation within the
ADP1111 when high peak current is required. To increase the
output current, an external PNP switch can be added (Figure
21). In this circuit, the ADP1111 provides base drive to Q1
through R3, while R4 ensures that Q1 turns off rapidly. Because
the ADP1111’s internal current limiting function will not work
in this circuit, R5 is provided for this purpose. With the value
shown, R5 limits current to 2 A. In addition to reducing power
dissipation on the ADP1111, this circuit also reduces the switch
voltage. When selecting an inductor value for the circuit of
Figure 21, the switch voltage can be calculated from the
formula:
V = V + V 0.6 V + 0.4 V 1 V
SW R5 Q1(SAT) ≅≅
I
LIM
V
IN
SW1
SW2
FB
GNDSETAO
ADP1111
NC
L1
D1
1N5821
R5
0.3Ω
1
INPUT
2
3
67 5 4
8
NC
C
INPUT
+
R1
R2
C
L
+
OUTPUT
R3
330Ω
R4
220ΩQ1
MJE210
Figure 21. High Current Step-Down Operation
REV. A
ADP1111
–11–
REV. 0
POSITIVE-TO-NEGATIVE CONVERSION
The ADP1111 can convert a positive input voltage to a negative
output voltage as shown in Figure 22. This circuit is essentially
identical to the step-down application of Figure 19, except that
the “output” side of the inductor is connected to power ground.
When the ADP1111’s internal power switch turns off, current
flowing in the inductor forces the output (–V
OUT
) to a negative
potential. The ADP1111 will continue to turn the switch on
until its FB pin is 1.25 V above its GND pin, so the output
voltage is determined by the formula:
V
OUT
=1. 25 V•1+R2
R1
⎛
⎝
⎜⎞
⎠
⎟
I
LIM
V
IN
SW1
SW2
FB
GNDSETAO
ADP1111
NC
L1
D1
1N5818
R
LIM
1
INPUT
2 3
67 5
4
8
NC
C
INPUT
+
R1
R2
C
L
+
OUTPUT
NEGATIVE
OUTPUT
Figure 22. Positive-to-Negative Converter
The design criteria for the step-down application also apply to
the positive-to-negative converter. The output voltage should be
limited to |6.2 V| unless a diode is inserted in series with the
SW2 pin (see Figure 20.) Also, D1 must again be a Schottky
diode to prevent excessive power dissipation in the ADP1111.
NEGATIVE-TO-POSITIVE CONVERSION
The circuit of Figure 23 converts a negative input voltage to a
positive output voltage. Operation of this circuit configuration is
similar to the step-up topology of Figure 18, except the current
through feedback resistor R2 is level-shifted below ground by a
PNP transistor. The voltage across R2 is V
OUT
–V
BEQ1
. How-
ever, diode D2 level-shifts the base of Q1 about 0.6 V below
ground thereby cancelling the V
BE
of Q1. The addition of D2
also reduces the circuit’s output voltage sensitivity to tempera-
ture, which otherwise would be dominated by the –2 mV V
BE
contribution of Q1. The output voltage for this circuit is
determined by the formula:
V
OUT
=1. 25 V•R2
R1
Unlike the positive step-up converter, the negative-to-positive
converter’s output voltage can be either higher or lower than the
input voltage.
I
LIM
V
IN
SW1
SW2
FB
GNDSETAO
ADP1111
NC
D1
1N5818
12
3
67 5 4
8
NC
C2
+
R1
10kΩ
C
L
+
POSITIVE
OUTPUT
R2
MJE210
R
LIM
NEGATIVE
INPUT
L1
D2
2N3906
Q1
Figure 23. ADP1111 Negative-to-Positive Converter
LIMITING THE SWITCH CURRENT
The ADP1111’s R
LIM
pin permits the switch current to be
limited with a single resistor. This current limiting action occurs
on a pulse by pulse basis. This feature allows the input voltage
to vary over a wide range without saturating the inductor or
exceeding the maximum switch rating. For example, a particular
design may require peak switch current of 800 mA with a 2.0 V
input. If V
IN
rises to 4 V, however, the switch current will
exceed 1.6 A. The ADP1111 limits switch current to 1.5 A and
thereby protects the switch, but the output ripple will increase.
Selecting the proper resistor will limit the switch current to
800 mA, even if V
IN
increases. The relationship between R
LIM
and maximum switch current is shown in Figure 6.
The I
LIM
feature is also valuable for controlling inductor current
when the ADP1111 goes into continuous-conduction mode.
Table I. Component Selection for Typical Converters
Input Output Output Circuit Inductor Inductor Capacitor
Voltage Voltage Current (mA) Figure Value Part No. Value Notes
2 to 3.1 5 90 mA 4 15 μH CD75-150K 33 μF*
2 to 3.1 5 10 mA 4 47 μH CTX50-1 10 μF
2 to 3.1 12 30 mA 4 15 μH CD75-150K 22 μF
2 to 3.1 12 10 mA 4 47 μH CTX50-1 10 μF
5 12 90 MA 4 33 μH CD75-330K 22 μF
51230mA 447μH CTX50-1 15 μF
6.5 to 11 5 50 mA 5 15 μH47μF**
12 to 20 5 300 mA 5 56 μH CTX50-4 47 μF**
20 to 30 5 300 mA 5 120 μH CTX100-4 47 μF**
5–57mA 656μH CTX50-4 47 μF
12 –5 250 mA 6 120 μH CTX100-4 100 μF**
NOTES
CD = Sumida.
CTX = Coiltronics.
**Add 47 Ω from I
LIM
to V
IN
.
**Add 220 Ω from I
LIM
to V
IN
.
REV. A
ADP1111
–12– REV. 0
This occurs in the step-up mode when the following condition is
met:
V
OUT
+V
DIODE
V
IN
−V
SW
<1
1−DC
where DC is the ADP1111’s duty cycle. When this relationship
exists, the inductor current does not go all the way to zero
during the time that the switch is OFF. When the switch turns
on for the next cycle, the inductor current begins to ramp up
from the residual level. If the switch ON time remains constant,
the inductor current will increase to a high level (see Figure 24).
This increases output ripple and can require a larger inductor
and capacitor. By controlling switch current with the I
LIM
resistor, output ripple current can be maintained at the design
values. Figure 25 illustrates the action of the I
LIM
circuit.
Figure 24.
Figure 25.
The internal structure of the I
LIM
circuit is shown in Figure 26.
Q1 is the ADP1111’s internal power switch that is paralleled by
sense transistor Q2. The relative sizes of Q1 and Q2 are scaled
so that I
Q2
is 0.5% of I
Q1
. Current flows to Q2 through an
internal 80 Ω resistor and through the R
LIM
resistor. These two
resistors parallel the base-emitter junction of the oscillator-
disable transistor, Q3. When the voltage across R1 and R
LIM
exceeds 0.6 V, Q3 turns on and terminates the output pulse. If
only the 80 Ω internal resistor is used (i.e. the I
LIM
pin is
connected directly to V
IN
), the maximum switch current will be
1.5 A. Figure 6 gives R
LIM
values for lower current-limit values.
72kHz
OSC
V
IN
POWER
SWITCH
SW2
SW1
R
LIM
DRIVER
80Ω
(INTERNAL)
I
LIM
I
Q1
200
V
IN
(EXTERNAL)
Q2
ADP1111
Q1
Q3
R1
Figure 26. ADP1111 Current Limit Operation
The delay through the current limiting circuit is approximately
1μs. If the switch ON time is reduced to less than 3 μs, accuracy
of the current trip-point is reduced. Attempting to program a
switch ON time of 1 μs or less will produce spurious responses
in the switch ON time; however, the ADP1111 will still provide
a properly regulated output voltage.
PROGRAMMING THE GAIN BLOCK
The gain block of the ADP1111 can be used as a low-battery
detector, error amplifier or linear post regulator. The gain block
consists of an op amp with PNP inputs and an open-collector
NPN output. The inverting input is internally connected to the
ADP1111’s 1.25 V reference, while the noninverting input is
available at the SET pin. The NPN output transistor will sink
about 300 μA.
Figure 27a shows the gain block configured as a low-battery
monitor. Resistors R1 and R2 should be set to high values to
reduce quiescent current, but not so high that bias current in
the SET input causes large errors. A value of 33 kΩ for R2 is a
good compromise. The value for R1 is then calculated from the
formula:
R1=V
LOBATT
−1. 25 V
1. 25 V
R2
where V
LOBATT
is the desired low battery trip point. Since the
gain block output is an open-collector NPN, a pull-up resistor
should be connected to the positive logic power supply.
ADP1111
1.25V
REF
GND
AO
5V
R
L
47k
TO
PROCESSOR
R1
R2
V
BAT
V
IN
SET
33k
R1= –––––––––
V
LB
–1.25V
35.1μA
V
LB
= BATTERY TRIP POINT
Figure 27a. Setting the Low Battery Detector Trip Point
200mA/div
200mA/div
REV. A
ADP1111
–13–
REV. 0
The circuit of Figure 27b may produce multiple pulses when
approaching the trip point due to noise coupled into the SET
input. To prevent multiple interrupts to the digital logic,
hysteresis can be added to the circuit (Figure 27). Resistor
RHYS, with a value of 1 MΩ to 10 MΩ, provides the hysteresis.
The addition of RHYS will change the trip point slightly, so the
new value for R1 will be:
R1=V
LOBATT
−1. 25 V
1. 25 V
R2
⎛
⎝
⎜⎞
⎠
⎟−V
L
−1. 25 V
R
L
+R
HYS
⎛
⎝
⎜⎞
⎠
⎟
where V
L
is the logic power supply voltage, R
L
is the pull-up
resistor, and R
HYS
creates the hysteresis.
ADP1111
1.25V
REF
GND
AO
5V
RL
47k
TO
PROCESSOR
R1
R2
VBAT
VIN
SET
RHYS
33k
1.6M
Figure 27b.
APPLICATION CIRCUITS
All Surface Mount 3 V to 5 V Step-Up Converter
This is the most basic application (along with the basic step-
down configuration to follow) of the ADP1111. It takes full
advantage of surface mount packaging for all the devices used in
the design. The circuit can provide +5 V at 100 mA of output
current and can be operated off of battery power for use in
portable equipment.
+
L1 D1
C
L
33μF
OUTPUT
R3*
(OPTIONAL)
MBRS120T3
20μH
CTX20-4 (5V @ 100mA)
INPUT +3V
I
LIM
V
IN
SW2
SW1
SENSE
GNDSETAO
ADP1111-5
NC
12
67 5
3
8
NC
4
Figure 28. All Surface Mount +3 V to +5 V Step-Up Converter
9 V to 5 V Step-Down Converter
This circuit uses a 9 V battery to generate a +5 V output. The
circuit will work down to 6.5 V, supplying 50 mA at this lower
limit. Switch current is limited to around 500 mA by the 100 Ω
resistor.
I
LIM
V
IN
SW1
SW2
SENSE
GNDSETAO
ADP1111-5
NC
L1
C
L
22μF
D1
1N5818
R
LIM
100Ω
1
+
15μH
CTX15-4
INPUT
2 3
67 5
4
8
NC
OUTPUT
(9V
IN
TO 5V @ 150mA,
6.5V
IN
TO 5V @ 50mA)
9V
Figure 29. 9 V to 5 V Step-Down Converter
20 V to 5 V Step-Down Converter
This circuit is similar to Figure 29, except it supplies much
higher output current and operates over a much wider range of
input voltage. As in the previous examples, switch current is
limited to 500 mA.
I
LIM
V
IN
SW1
SW2
SENSE
GNDSETAO
ADP1111-5
NC
L1
C
L
47μF
D1
1N5818
R
LIM
100Ω
1
+
68μH
CTX68-4
12V TO 28V
INPUT
2 3
67 5
4
8
NC
OUTPUT
(+5V @ 300mA)
Figure 30. 20 V to 5 V Step-Down Converter
+5 V to –5 V Converter
This circuit is essentially identical to Figure 22, except it uses a
fixed-output version of the ADP1111 to simplify the design
somewhat.
I
LIM
V
IN
SW1
SW2
GNDSETAO
ADP1111-5
NC
L1
C
L
33μF
D1
1N5818
R
LIM
100Ω
1
+
33μH
CTX33-2
12V TO 28V
INPUT
2 3
67 5
4
8
NC –5V
@ 75mA
SENSE
Figure 31. +5 V to –5 V Converter
REV. A
I
LIM
V
IN
SW1
SW2
GNDSETAO
ADP1111-5
NC
L1
C
L
33μF
D1
1N5818
R
LIM
100Ω
1
+
33μH
CTX33-2
5V TO 25V
INPUT
2 3
67 5
4
8
NC –5V
@ 75mA
SENSE
ADP1111
–14– REV. 0
Voltage-Controlled Positive-to-Negative Converter
By including an op amp in the feedback path, a simple positive-
to-negative converter can be made to give an output that is a
linear multiple of a controlling voltage, Vc. The op amp, an
OP196, rail-to-rail input and output amplifier, sums the
currents from the output and controlling voltage and drives the
FB pin either high or low, thereby controlling the on-board
oscillator. The 0.22 Ω resistor limits the short-circuit current to
about 3 A and, along with the BAT54 Schottky diode, helps
limit the peak switch current over varying input voltages. The
external power switch features an active pull-up to speed up the
turn-off time of the switch. Although an IRF9530 was used in
the evaluation, almost any device that can handle at least 3 A of
peak current at a VDS of at least 50 V is suitable for use in this
application, provided that adequate attention is paid to power
dissipation. The circuit can deliver 2 W of output power with a
+6-volt input from a control voltage range of 0 V to 5 V.
I
LIM
V
IN
SW1
SW2
FB
GNDSETAO
ADP1111
NC
21
3
67 5 4
8
NC
L1
C
L
47μF
35V
OUTPUT
R
LIM
INPUT
+
+5V TO +12V
0.22Ω
2kΩ
3
4
672
1kΩ200kΩ
39kΩ
V
IN
1N4148
IRF9530
20μH
D1
IN5821
–V
OUT
= –5.13 *V
C
2W MAXIMUM OUTPUT
1N5231
CTX20-4
V
C
(0V TO +5V)
2N3904
51Ω
BAT54
Figure 32. Voltage Controlled Positive-to-Negative
Converter
+3 V to –22 V LCD Bias Generator
This circuit uses an adjustable-output version of the ADP1111
to generate a +22.5 V reference output that is level-shifted to
give an output of –22 V. If operation from a +5 volt supply is
desired, change R1 to 47 ohms. The circuit will deliver 7 mA
with a 3 volt supply and 40 mA with a 5 volt supply.
L1
CL
0.1μF
OUTPUT
25μH
D1
1N4148
1N5818
1N5818
732kΩ
42.2kΩ
22μF
4.7
μF
+3V
2xAA
CELLS
RLIM
100Ω
–22V OUTPUT
7mA @ 2V INPUT
L1 = CTX25-4
ILIM VIN SW1
SW2
FB
GNDSETAO
ADP1111
NC
12
3
67 5 4
8
NC +
+
Figure 33. 3 V to –22 V LCD Bias Generator
High Power, Low Quiescent Current Step-Down Converter
By making use of the fact that the feedback pin directly controls
the internal oscillator, this circuit achieves a shutdown-like state
by forcing the feedback pin above the 1.25 V comparator
threshold. The logic level at the 1N4148 diode anode needs to
be at least 2 V for reliable standby operation.
The external switch driver circuit features an active pull-up
device, a 2N3904 transistor, to ensure that the power MOSFET
turns off quickly. Almost any power MOSFET will do as the
switch as long as the device can withstand the 18 volt V
GS
and is
reasonably robust. The 0.22 Ω resistor limits the short-circuit
current to about 3 A and, along with the BAT54 Schottky
diode, helps to limit the peak switch current over varying input
voltages.
C
L
220μF
+
IRF9540
D1
IN5821
NC
R
LIM
INPUT
+8V TO +18V
0.22
2k
1N4148
BAT54
2N3904
G
SD
121kΩ
40.2kΩ
51Ω
1N4148
+5V
500mA
OPERATE/STANDBY
2V ≤ V
IN
≤ 5
I
LIM
V
IN
SW1
SW2
FB
GNDSETAO
ADP1111
1
2
3
67 5 4
8
NC
LI
20μH
LI = COILTRONICS CTX20-4
Figure 34. High Power, Low Quiescent Current Step-Down
Converter
NOTES
1. All inductors referenced are Coiltronics CTX-series except
where noted.
2. If the source of power is more than an inch or so from the
converter, the input to the converter should be bypassed with
approximately 10 μF of capacitance. This capacitor should
be a good quality tantalum or aluminum electrolytic.
REV. A
ADP1111
REV.A ‐15‐
OUTLINE DIMENSIONS
COMPLIANT TO JEDEC STANDARDS MS-001
CONT ROLLI NG DIMENSIONS ARE IN I NCHES ; MILLIMET E R DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESI GN.
CORNER LEADS MAY BE CONFIGURED AS WHO LE OR HALF LEADS.
070606-A
0.022 (0.56)
0.018 (0.46)
0.014 (0.36)
SEATING
PLANE
0.015
(0.38)
MIN
0.210 (5.33)
MAX
0.150 (3. 81)
0.130 (3. 30)
0.115 (2.92)
0.070 (1.78)
0.060 (1.52)
0.045 (1.14)
8
14
5
0.280 (7.11)
0.250 (6.35)
0.240 (6.10)
0.100 (2.54)
BSC
0.400 (10.16)
0.365 (9.27)
0.355 (9.02)
0.060 (1.52)
MAX
0.430 (10.92)
MAX
0.014 (0.36)
0.010 (0.25)
0.008 (0.20)
0.325 (8.26)
0.310 (7.87)
0.300 (7.62)
0.195 (4.95)
0.130 (3.30)
0.115 (2.92)
0.015 (0.38)
GAUGE
PLANE
0.005 (0.13)
MIN
Figure 35. 8-Lead Plastic Dual In-Line Package (PDIP)
Narrow Body (N-8)
Dimensions shown in inches and (millimeters)
CONTROLLING DIMENS IONS ARE IN MI LLI METE R S ; I N CH DIMENSIONS
(I N PAR E NTHESES) AR E ROUNDED- OFF M ILLIM E TER EQUIVALENT S FOR
REF E RENCE O N LY AND ARE NO T AP P R OPRI ATE FOR US E IN DESIGN.
COM P LIANT TO JEDEC STANDARDS M S - 01 2- AA
012407-A
0.25 (0.0098)
0.17 (0.0067)
1.27 (0. 0500)
0.40 (0. 0157)
0.50 (0. 0196)
0.25 (0. 0099) 45°
8°
0°
1.75 (0.0688)
1.35 (0.0532)
SEATING
PLANE
0.2 5 ( 0.0098)
0.1 0 ( 0.0040)
4
1
85
5.00 (0.1968)
4.80 (0.1890)
4.00 ( 0.1574)
3.80 ( 0.1497)
1.27 (0.0500)
BSC
6.20 (0. 2441)
5.80 (0. 2284)
0.51 (0.0201)
0.31 (0.0122)
COPLANARITY
0.10
Figure 36. 8-Lead Standard Small Outline Package (SOIC_N)
Narrow Body (R-8)
Dimensions shown in millimeters and (inches)
ORDERING GUIDE
Model1
Output
Voltage Temperature Range Package Description
Package
Option
ADP1111ANZ-12 12 V −40°C to +85°C 8-Lead Plastic Dual In-Line Package [PDIP] N-8
ADP1111ANZ-3.3 3.3 V −40°C to +85°C 8-Lead Plastic Dual In-Line Package [PDIP] N-8
ADP1111ANZ-5 5 V −40°C to +85°C 8-Lead Plastic Dual In-Line Package [PDIP] N-8
ADP1111ARZ ADJ −40°C to +85°C 8-Lead Standard Small Outline Package [SOIC_N] R-8
ADP1111ARZ-12 12 V −40°C to +85°C 8-Lead Standard Small Outline Package [SOIC_N] R-8
ADP1111ARZ-12-REEL 12 V −40°C to +85°C 8-Lead Standard Small Outline Package [SOIC_N] R-8
ADP1111ARZ-3.3 3.3 V −40°C to +85°C 8-Lead Standard Small Outline Package [SOIC_N] R-8
ADP1111ARZ-5 5 V −40°C to +85°C 8-Lead Standard Small Outline Package [SOIC_N] R-8
ADP1111ARZ-5-REEL 5 V −40°C to +85°C 8-Lead Standard Small Outline Package [SOIC_N] R-8
ADP1111ARZ-REEL ADJ −40°C to +85°C 8-Lead Standard Small Outline Package [SOIC_N] R-8
1 Z = RoHS Compliant Part.
©1996–2009 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D08365-0-11/09(A)
